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Data Sheet No. PD94173
IRU3037 / IRU3037A
8-PIN SYNCHRONOUS PWM CONTROLLER FEATURES
Synchronous Controller in 8-Pin Package Operating with single 5V or 12V supply voltage Internal 200KHz Oscillator (400KHz for IRU3037A) Soft-Start Function Fixed Frequency Voltage Mode 500mA Peak Output Drive Capability Protects the output when control FET is shorted
DESCRIPTION
The IRU3037 controller IC is designed to provide a low cost synchronous Buck regulator for on-board DC to DC converter applications. With the migration of today's ASIC products requiring low supply voltages such as 1.8V and lower, together with currents in excess of 3A, traditional linear regulators are simply too lossy to be used when input supply is 5V or even in some cases with 3.3V input supply. The IRU3037 together with dual N-channel MOSFETs such as IRF7313, provide a low cost solution for such applications. This device features an internal 200KHz oscillator (400KHz for "A" version), under-voltage lockout for both Vcc and Vc supplies, an external programmable soft-start function as well as output under-voltage detection that latches off the device when an output short is detected.
APPLICATIONS
DDR memory source sink Vtt application Low cost on-board DC to DC such as 5V to 3.3V, 2.5V or 1.8V Graphic Card Hard Disk Drive
TYPICAL APPLICATION
5V 12V C3 0.1uF C4 1uF C2 10TPB100M, 100uF, 55mV L1 1uH C1 47uF
Vcc
Vc HDrv Q1 1/2 of IRF7313 L2 1.5V/5A C7 2 X 6TPC150M, 150uF, 40mV
C8 0.1uF
SS
U1 IRU3037
LDrv
D05022P-562, 5.6uH, 5.3A Q2 1/2 of IRF7313
C9 2200pF R4 24k
Comp Fb Gnd R5 1.24K, 1%
R3 249, 1%
Figure 1 - Typical application of IRU3037 or IRU3037A.
PACKAGE ORDER INFORMATION
TA (C) 0 To 70 0 To 70 0 To 70 0 To 70 DEVICE IRU3037CS IRU3037CF IRU3037ACS IRU3037ACF PACKAGE 8-Pin Plastic SOIC 8-Pin Plastic TSSOP 8-Pin Plastic SOIC 8-Pin Plastic TSSOP FREQUENCY 200KHz 200KHz 400KHz 400KHz
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IRU3037 / IRU3037A
ABSOLUTE MAXIMUM RATINGS
Vcc Supply Voltage .................................................. Vc Supply Voltage .................................................... Storage Temperature Range ...................................... Operating Junction Temperature Range ..................... 30V 30V -65C To 150C 0C To 125C
PACKAGE INFORMATION
8-PIN PLASTIC SOIC (S) TOP VIEW 8-PIN PLASTIC TSSOP (F) TOP VIEW
Fb 1 Vcc 2 LDrv 3 Gnd 4
JA=160C/W
8 SS 7 Comp 6 Vc 5 HDrv
Fb 1 Vcc 2 LDrv 3 Gnd 4
8 SS 7 Comp 6 Vc 5 HDrv
JA=124C/W
ELECTRICAL SPECIFICATIONS
Unless otherwise specified, these specifications apply over Vcc=5V, Vc=12V and TA=0 to 70C. Typical values refer to TA=25C. Low duty cycle pulse testing is used which keeps junction and case temperatures equal to the ambient temperature. PARAMETER Reference Voltage Fb Voltage Fb Voltage Line Regulation UVLO UVLO Threshold - Vcc UVLO Hysteresis - Vcc UVLO Threshold - Vc UVLO Hysteresis - Vc UVLO Threshold - Fb UVLO Hysteresis - Fb Supply Current Vcc Dynamic Supply Current Vc Dynamic Supply Current Vcc Static Supply Current Vc Static Supply Current Soft-Start Section Charge Current SYM
VFB LREG
TEST CONDITION IRU3037 IRU3037A 5MIN 1.225 0.784
TYP 1.250 0.800 0.2 4.2 0.25 3.3 0.2 0.6 0.4 0.1 5 7 3.3 1 -20
MAX 1.275 0.816 0.35 4.4 3.5 0.8 0.5
UNITS
V % V V V V V V
UVLO Vcc Supply Ramping Up UVLO Vc Supply Ramping Up UVLO Fb Fb Ramping Down (IRU3037)
4.0 3.1 0.4 0.3
(IRU3037A)
Dyn Icc Dyn Ic ICCQ ICQ SSIB
Freq=200KHz, CL=1500pF Freq=200KHz, CL=1500pF SS=0V SS=0V SS=0V
2 2 1 0.5 -10
8 10 6 4.5 -30
mA mA mA mA mA
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IRU3037 / IRU3037A
PARAMETER Error Amp Fb Voltage Input Bias Current Fb Voltage Input Bias Current Transconductance Oscillator Frequency Ramp-Amplitude Voltage Output Drivers Rise Time Fall Time Dead Band Time Max Duty Cycle Min Duty Cycle SYM
IFB1 IFB2 GM Freq VRAMP Tr Tf TDB TON TOFF
TEST CONDITION SS=3V, Fb=1V SS=0V, Fb=1V
MIN
TYP -0.1 -64 600 200 400 1.25 50 50 150 90 0
MAX
UNITS mA mA mmho KHz V ns ns ns % %
450 IRU3037 IRU3037A 180 360 1.225
750 220 440 1.275 100 100 250 95
CL=1500pF CL=1500pF Fb=1V, Freq=200KHz Fb=1.5V 50 85 0
PIN DESCRIPTIONS
PIN# 1 PIN SYMBOL PIN DESCRIPTION Fb This pin is connected directly to the output of the switching regulator via resistor divider to provide feedback to the Error amplifier. Vcc This pin provides biasing for the internal blocks of the IC as well as power for the low side driver. A minimum of 1mF, high frequency capacitor must be connected from this pin to ground to provide peak drive current capability. Output driver for the synchronous power MOSFET. This pin serves as the ground pin and must be connected directly to the ground plane. A high frequency capacitor (0.1 to 1mF) must be connected from V5 and V12 pins to this pin for noise free operation. Output driver for the high side power MOSFET. This pin is connected to a voltage that must be at least 4V higher than the bus voltage of the switcher (assuming 5V threshold MOSFET) and powers the high side output driver. A minimum of 1mF, high frequency capacitor must be connected from this pin to ground to provide peak drive current capability. Compensation pin of the error amplifier. An external resistor and capacitor network is typically connected from this pin to ground to provide loop compensation. This pin provides soft-start for the switching regulator. An internal current source charges an external capacitor that is connected from this pin to ground which ramps up the output of the switching regulator, preventing it from overshooting as well as limiting the input current. The converter can be shutdown by pulling this pin below 0.5V.
2
3 4
LDrv Gnd
5 6
HDrv Vc
7
Comp
8
SS
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IRU3037 / IRU3037A
BLOCK DIAGRAM
Vcc 0.2V Bias Generator 3V 1.25V
4.0V 3V Vc 20uA 3.5V SS 8 64uA Max Oscillator Ct S 1.25V 25K POR Error Comp Error Amp R 25K FbLo Comp Reset Dom Q 0.2V
POR
6 Vc
5 HDrv
2 Vcc
Fb 1
3 LDrv 0.5V
Comp 7 POR
4 Gnd
Figure 2 - Simplified block diagram of the IRU3037.
THEORY OF OPERATION
Introduction The IRU3037 is a fixed frequency, voltage mode synchronous controller and consists of a precision reference voltage, an error amplifier, an internal oscillator, a PWM comparator, 0.5A peak gate driver, soft-start and shutdown circuits (See the block diagram). The output voltage of the synchronous converter is set and controlled by the output of the error amplifier; this is the amplified error signal from the sensed output voltage and the reference voltage. This voltage is compared to a fixed frequency linear sawtooth ramp and generates fixed frequency pulses of variable duty-cycle, which drives the two N-channel external MOSFETs. The timing of the IC is provided through an internal oscillator circuit which uses on-chip capacitor to set the oscillation frequency to 200 KHz (400 KHz for "A" version). Soft-Start The IRU3037 has a programmable soft-start to control the output voltage rise and limit the current surge at the start-up. To ensure correct start-up, the soft-start sequence initiates when the Vc and Vcc rise above their threshold (3.3V and 4.2V respectively) and generates the Power On Reset (POR) signal. Soft-start function operates by sourcing an internal current to charge an external capacitor to about 3V. Initially, the soft-start function clamps the E/A's output of the PWM converter. As the charging voltage of the external capacitor ramps up, the PWM signals increase from zero to the point the feedback loop takes control. Short-Circuit Protection The outputs are protected against the short-circuit. The IRU3037 protects the circuit for shorted output by sensing the output voltage (through the external resistor divider). The IRU3037 shuts down the PWM signals, when the output voltage drops below 0.6V (0.4V for IRU3037A). The IRU3037 also protects the output from over-voltaging when the control FET is shorted. This is done by turning on the sync FET with the maximum duty cycle. Under-Voltage Lockout The under-voltage lockout circuit assures that the MOSFET driver outputs remain in the off state whenever the supply voltage drops below set parameters. Lockout occurs if Vc and Vcc fall below 3.3V and 4.2V respectively. Normal operation resumes once Vc and Vcc rise above the set values.
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IRU3037 / IRU3037A
APPLICATION INFORMATION
Design Example: The following example is a typical application for IRU3037, the schematic is Figure 18 on page 14. VIN = 5V VOUT = 3.3V IOUT = 4A DVOUT = 100mV fS = 200KHz Output Voltage Programming Output voltage is programmed by reference voltage and external voltage divider. The Fb pin is the inverting input of the error amplifier, which is internally referenced to 1.25V (0.8V for IRU3037A). The divider is ratioed to provide 1.25V at the Fb pin when the output is at its desired value. The output voltage is defined by using the following equation: VOUT = VREF 3 Soft-Start Programming The soft-start timing can be programmed by selecting the soft start capacitance value. The start up time of the converter can be calculated by using:
tSTART = 75 3 Css (ms)
(2)
Where: CSS is the soft-start capacitor (mF) For a start-up time of 7.5ms, the soft-start capacitor will be 0.1mF. Choose a ceramic capacitor at 0.1mF. Shutdown The converter can be shutdown by pulling the soft-start pin below 0.5V. The control MOSFET turns off and the synchronous MOSFET turns on during shutdown. Boost Supply Vc To drive the high-side switch it is necessary to supply a gate voltage at least 4V greater than the bus voltage. This is achieved by using a charge pump configuration as shown in Figure 18. The capacitor is charged up to approximately twice the bus voltage. A capacitor in the range of 0.1mF to 1mF is generally adequate for most applications. In application, when a separate voltage source is available the boost circuit can be avoided as shown in Figure 1. Input Capacitor Selection The input filter capacitor should be based on how much ripple the supply can tolerate on the DC input line. The larger capacitor, the less ripple expected but consider should be taken for the higher surge current during the power-up. The IRU3037 provides the soft-start function which controls and limits the current surge. The value of the input capacitor can be calculated by the following formula: CIN = IIN 3 Dt DV (3)
(
R6 1+ R5
)
(1)
When an external resistor divider is connected to the output as shown in Figure 3.
VOUT IRU3037
Fb R5 R6
Figure 3 - Typical application of the IRU3037 for programming the output voltage. Equation (1) can be rewritten as: R6 = R5 3
( VOUT - 1) VREF
Choose R5 = 1KV will result to: R6 = 1.65KV If the high value feedback resistors are used, the input bias current of the Fb pin could cause a slight increase in output voltage. The output voltage set point can be more accurate by using precision resistor.
Where: CIN is the input capacitance (mF) IIN is the input current (A) Dt is the turn on time of the high-side switch (ms) DV is the allowable peak to peak voltage ripple (V)
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IRU3037 / IRU3037A
Assuming the following: DV = 1%(VIN), Efficiency(h) = 90% 1 fS VO 3 IO IIN = h 3 VIN Dt = D 3 Y Dt = 3.3ms IIN = 2.93A Y
VIN - VOUT = L L = (VIN - V ) OUT Di VOUT 1 Dt = D fS ; D = VIN Dt ; VOUT VIN Di fS
(5)
By using equation (3), CIN = 193.3mF For higher efficiency, low ESR capacitor is recommended. Choose two 100mF capacitors. The Sanyo TPB series PosCap capacitor 100mF, 10V with 55mV ESR is a good choice. Output Capacitor Selection The criteria to select the output capacitor is normally based on the value of the Effective Series Resistance (ESR). In general, the output capacitor must have low enough ESR to meet output ripple and load transient requirements, yet have high enough ESR to satisfy stability requirements. The ESR of the output capacitor is calculated by the following relationship: DVO (4) ESR [ DIO Where: DVO = Output Voltage Ripple DIO = Output Current DVO=100mV and DIO=4A, result to ESR=25mV The Sanyo TPC series, PosCap capacitor is a good choice. The 6TPC150M 150mF, 6.3V has an ESR 40mV. Selecting two of these capacitors in parallel, results to an ESR of 20mV which achieves our low ESR goal. The capacitor value must be high enough to absorb the inductor's ripple current. The larger the value of capacitor, the lower will be the output ripple voltage. Inductor Selection The inductor is selected based on output power, operating frequency and efficiency requirements. Low inductor value causes large ripple current, resulting in the smaller size, but poor efficiency and high output noise. Generally, the selection of inductor value can be reduced to desired maximum ripple current in the inductor (i). The optimum point is usually found between 20% and 50% ripple of the output current. For the buck converter, the inductor value for desired operating ripple current can be determined using the following relation:
Where: VIN = Maximum Input Voltage VOUT = Output Voltage i = Inductor Ripple Current fS = Switching Frequency t = Turn On Time D = Duty Cycle If Di = 20%(IO), then the output inductor will be: L = 7mH The Toko D124C series provides a range of inductors in different values, low profile suitable for large currents, 10mH, 4.2A is a good choice for this application. This will result to a ripple approximately 14% of output current. Power MOSFET Selection The IRU3037 uses two N-Channel MOSFETs. The selections criteria to meet power transfer requirements is based on maximum drain-source voltage (VDSS), gatesource drive voltage (VGS), maximum output current, Onresistance RDS(ON) and thermal management. The MOSFET must have a maximum operating voltage (VDSS) exceeding the maximum input voltage (VIN). The gate drive requirement is almost the same for both MOSFETs. Logic-level transistor can be used and caution should be taken with devices at very low VGS to prevent undesired turn-on of the complementary MOSFET, which results a shoot-through current. The total power dissipation for MOSFETs includes conduction and switching losses. For the Buck converter the average inductor current is equal to the DC load current. The conduction loss is defined as: PCOND (Upper Switch) = ILOAD 3 RDS(ON) 3 D 3 q PCOND (Lower Switch) = ILOAD 3 RDS(ON) 3 (1 - D) 3 q q = RDS(ON) Temperature Dependency The RDS(ON) temperature dependency should be considered for the worst case operation. This is typically given in the MOSFET data sheet. Ensure that the conduction losses and switching losses do not exceed the package ratings or violate the overall thermal budget.
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IRU3037 / IRU3037A
For this design, IRF7301 is a good choice. The device provides low on-resistance in a compact SOIC 8-Pin package. The IRF7301 has the following data: VDSS = 20V ID = 5.2A RDS(ON) = 0.05V The total conduction losses will be:
PCON(TOTAL)=PCON(Upper Switch)+PCON(Lower Switch)
These values are taken under a certain condition test. For more detail please refer to the IRF7301 data sheet. By using equation (6), we can calculate the switching losses. PSW = 0.186W Feedback Compensation The IRU3037 is a voltage mode controller; the control loop is a single voltage feedback path including error amplifier and error comparator. To achieve fast transient response and accurate output regulation, a compensation circuit is necessary. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency and adequate phase margin (greater than 458). The output LC filter introduces a double pole, -40dB/ decade gain slope above its corner resonant frequency, and a total phase lag of 1808 (see Figure 5). The Resonant frequency of the LC filter expressed as follows:
FLC = 2p 1 Lo Co
PCON(TOTAL) = ILOAD 3 RDS(ON) 3 q q = 1.5 according to the IRF7301 data sheet for 1508C junction temperature PCON(TOTAL) = 1.2W The switching loss is more difficult to calculate, even though the switching transition is well understood. The reason is the effect of the parasitic components and switching times during the switching procedures such as turn-on / turnoff delays and rise and fall times. With a linear approximation, the total switching loss can be expressed as: VDS(OFF) tr + tf 3 ILOAD PSW = 3 (6) 2 T Where: VDS(OFF) = Drain to Source Voltage at off time tr = Rise Time tf = Fall Time T = Switching Period ILOAD = Load Current The switching time waveform is shown in figure 4.
VDS 90%
2
(7)
Figure 5 shows gain and phase of the LC filter. Since we already have 1808 phase shift just from the output filter, the system risks being unstable.
Gain 0dB -40dB/decade Phase 08
-1808 FLC Frequency
FLC
Frequency
Figure 5 - Gain and phase of LC filter. The IRU3037's error amplifier is a differential-input transconductance amplifier. The output is available for DC gain control or AC phase compensation.
td(ON) tr td(OFF) tf
10% VGS
Figure 4 - Switching time waveforms. From IRF7301 data sheet we obtain:
tr = 42ns tf = 51ns
The E/A can be compensated with or without the use of local feedback. When operated without local feedback the transconductance properties of the E/A become evident and can be used to cancel one of the output filter poles. This will be accomplished with a series RC circuit from Comp pin to ground as shown in Figure 6.
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IRU3037 / IRU3037A
Note that this method requires that the output capacitor should have enough ESR to satisfy stability requirements. In general the output capacitor's ESR generates a zero typically at 5KHz to 50KHz which is essential for an acceptable phase margin. The ESR zero of the output capacitor expressed as follows: 1 FESR = (8) 2p 3 ESR 3 Co
VOUT R6 Fb
Where: VIN = Maximum Input Voltage VOSC = Oscillator Ramp Voltage Fo = Crossover Frequency FESR = Zero Frequency of the Output Capacitor FLC = Resonant Frequency of the Output Filter R5 and R6 = Resistor Dividers for Output Voltage Programming gm = Error Amplifier Gain For: VIN = 5V VOSC = 1.25V Fo = 30KHz FESR = 26.52KHz FLC = 2.9KHz R5 = 1K R6 = 1.65K gm = 600mhmo This results to R4=104.4KV. Choose R4=105KV To cancel one of the LC filter poles, place the zero before the LC filter resonant frequency pole: FZ 75%FLC
R5 VREF
Gain(dB)
E/A
Comp V e C9 R4
H(s) dB
FZ
Frequency
Figure 6 - Compensation network without local feedback and its asymptotic gain plot. The transfer function (Ve / VOUT) is given by: H(s) = gm 3
FZ 0.75 3
1 2p LO 3 CO
(13)
(
R5 R6 + R5
SR4C9 ) 3 1 +SC9
(9)
For: Lo = 10mH Co = 300mF FZ = 2.17KHz R4 = 86.6KV Using equations (11) and (13) to calculate C9, we get:
The (s) indicates that the transfer function varies as a function of frequency. This configuration introduces a gain and zero, expressed by: R5 |H(s)| = gm 3 3 R4 (10) R6 3 R5 FZ = 1 2p 3 R4 3 C9 (11)
C9 = 698pF Choose: C9 = 680pF One more capacitor is sometimes added in parallel with C9 and R4. This introduces one more pole which is mainly used to supress the switching noise. The additional pole is given by: 1 FP = C9 3 CPOLE 2p 3 R4 3 C9 + CPOLE The pole sets to one half of switching frequency which results in the capacitor CPOLE: CPOLE = 1 1 p 3 R4 3 fS C9 fS for FP << 2 1 p 3 R4 3 fS
The gain is determined by the voltage divider and E/A's transconductance gain. First select the desired zero-crossover frequency (Fo): Fo > FESR and FO [ (1/5 ~ 1/10)3 fS Use the following equation to calculate R4: R4 = 1 VOSC Fo3FESR R5 + R6 3 3 3 VIN FLC2 R5 gm (12)
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IRU3037 / IRU3037A
For a general solution for unconditionally stability for any type of output capacitors, in a wide range of ESR values we should implement local feedback with a compensation network. The typically used compensation network for voltage-mode controller is shown in Figure 7.
ZIN C10 R8 R6 Fb R5 VREF VOUT C12 R7 C11 Zf
FP1 = 0 FP2 = FP3 = 1 2p3R83C10 1 C123C11 2p3R73 C12+C11 1 2p3R73C11
(
)
1 2p3R73C12
FZ1 =
1 1 FZ2 = 2p3C103(R6 + R8) 2p3C103R6 Cross Over Frequency: VIN 1 FO = R73C103 3 VOSC 2p3Lo3Co Where: VIN = Maximum Input Voltage VOSC = Oscillator Ramp Voltage Lo = Output Inductor Co = Total Output Capacitors
E/A
Comp
V e
(15)
Gain(dB)
H(s) dB
FZ1
FZ2
FP2
FP3
Frequency
Figure 7 - Compensation network with local feedback and its asymptotic gain plot. In such configuration, the transfer function is given by: 1 - gmZf Ve = VOUT 1 + gmZIN The error amplifier gain is independent of the transconductance under the following condition:
The stability requirement will be satisfied by placing the poles and zeros of the compensation network according to following design rules. The consideration has been taken to satisfy condition (14) regarding transconductance error amplifier. 1) Select the crossover frequency: Fo < FESR and Fo [ (1/10 ~ 1/6)3 fS 2) Select R7, so that R7 >> 2
gmZf >> 1
and
gmZIN >>1
(14)
gm
By replacing ZIN and Zf according to Figure 7, the transformer function can be expressed as: H(s)= 1 3 SR6(C12+C11)
(1+SR7C11)3[1+SC10(R6+R8)] C123C11 1+SR7 3(1+SR8C10) 4) Place third pole at the half of the switching frequency. C12+C11 Fs FP3 = 2 As known, transconductance amplifier has high imped1 C12 = ance (current source) output, therefore, consider should 2p 3 R7 3 FP3 be taken when loading the E/A output. It may exceed its source/sink output current capability, so that the ampliC12 > 50pF fier will not be able to swing its output voltage over the If not, change R7 selection. necessary range. 5) Place R7 in (15) and calculate C10: The compensation network has three poles and two ze2p 3 Lo 3 Fo 3 Co VOSC ros and they are expressed as follows: C10 [ 3 R7 VIN
3) Place first zero before LC's resonant frequency pole. FZ1 75% FLC 1 C11 = 2p 3 FZ1 3 R7
[
(
)]
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IRU3037 / IRU3037A
6) Place second pole at the ESR zero. FP2 = FESR 1 R8 = 2p 3 C10 3 FP2 Check if R8 > 1 Start to place the power components, make all the connection in the top layer with wide, copper filled areas. The inductor, output capacitor and the MOSFET should be close to each other as possible. This helps to reduce the EMI radiated by the power traces due to the high switching currents through them. Place input capacitor directly to the drain of the high-side MOSFET, to reduce the ESR replace the single input capacitor with two parallel units. The feedback part of the system should be kept away from the inductor and other noise sources, and be placed close to the IC. In multilayer PCB use one layer as power ground plane and have a control circuit ground (analog ground), to which all signals are referenced. The goal is to localize the high current path to a separate loop that does not interfere with the more sensitive analog control function. These two grounds must be connected together on the PC board layout at a single point. Figure 8 shows a suggested layout for the critical components, based on the schematic on page 14.
PGnd
C1 C2A, B
gm
If R8 is too small, increase R7 and start from step 2. 7) Place second zero around the resonant frequency. FZ2 = FLC 1 R6 = - R8 2p 3 C10 3 FZ2 8) Use equation (1) to calculate R5. R5 = VREF 3 R6 VOUT - VREF
These design rules will give a crossover frequency approximately one-tenth of the switching frequency. The higher the band width, the potentially faster the load transient speed. The gain margin will be large enough to provide high DC-regulation accuracy (typically -5dB to 12dB). The phase margin should be greater than 458 for overall stability. IC Quiescent Power Dissipation Power dissipation for IC controller is a function of applied voltage, gate driver loads and switching frequency. The IC's maximum power dissipation occurs when the IC operating with single 12V supply voltage (Vcc=12V and Vc24V) at 400KHz switching frequency and maximum gate loads. Figures 9 and 10 show voltage vs. current, when the gate drivers loaded with 470pF, 1150pF and 1540pF capacitors. The IC's power dissipation results to an excessive temperature rise. This should be considered when using IRU3037A for such application. Layout Consideration The layout is very important when designing high frequency switching converters. Layout will affect noise pickup and can cause a good design to perform with less than expected results.
PGnd PGnd
C7A, B
L1 Vin
8 7 6 5
Vout
L2 Q1
1 2 3 4
DDDC 3215
5
4
C4 6
U1 3 IRU3037
2 C3 R5 1 R6
Single Point Analog Gnd Connect to Power Ground plane
R4 C9 7
Analog Gnd
C8 8
Analog Gnd
Figure 8 - Suggested layout. (Topside shown only)
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IRU3037 / IRU3037A
TYPICAL PERFORMANCE CHARACTERISTICS
IRU3037A Vcc vs. Icc
@470PF, 1150PF and 1540PF Gate Load
TA = 258C
14 12 10 8 6 4 2 0 0 2 4 6 Vcc (V) 8
CLOAD =1540pF CLOAD =1150pF
Icc (mA)
CLOAD =470pF
10
12
14
Figure 9
IRU3037A Vc vs. Ic
@470PF, 1150PF and 1540PF Gate Load
TA = 258C
30 25 Ic (ma) 20 15 10 5 0 0 2 4 6 8 10 12 14 16 18 20 22 24 26 Vc (V)
Figure 10
CLOAD =470pF CLOAD =1540pF CLOAD =1150pF
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IRU3037 / IRU3037A
TYPICAL PERFORMANCE CHARACTERISTICS
IRU3037 Output Voltage
1.3 240
IRU3037 Output Frequency
230
1.28 220 Max
Max
210 1.26 Kilo Hertz 1.24 190 1.22 1.2 -40C 0C Output Voltage +50C +100C Spec Min. +150C
Volts
200
180
170
160 -40C 0C +50C +100C Spec Max. Spec Min. +150C
Min
Spec Max. Oscillation Frequency
Min
Figure 11
Figure 12
IRU3037 Maximum Duty Cycle
92.0%
90.0%
88.0% Percent Duty Cycle
86.0%
84.0%
82.0%
80.0% -40C -25C 0C +25C +50C +75C +100C +125C +150C
Max Duty Cycle
Figure 13
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IRU3037 / IRU3037A
TYPICAL PERFORMANCE CHARACTERISTICS
IRU3037A Output Voltage
820 460
IRU3037A Output Frequency
Max 440 810 Max
420
800 400
milli Volts
790
Kilo Hertz
380
Min
780
360
Min
340
770 320
760 -40C -25C 0C +25C +50C Spec Max. +75C +100C +150C
300 -40C -25C 0C +25C +50C +75C +100C +150C
Output Voltage
Spec Min.
Oscillation Frequency
Spec Max.
Spec Min.
Figure 14
IRU3037 / IRU3037A Transconductance ( GM )
Figure 15
IRU3037 / IRU3037A Rise Time / Fall Time CL = 1500pF
1000 50 900 45 800 40 700 35 600 nano Seconds micro Mho's
30
500
25
400
20
300
15
200
10
100
5
0 -40C -25C 0C +25C +50C Negative load GM +75C +100C
0 -40C -25C 0C +25C Rise Time +50C Fall time +75C +100C
Positive load GM
Figure 16
Figure 17
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IRU3037 / IRU3037A TYPICAL APPLICATION
Single Supply 5V Input
5V D1 1N4148 D3 1N4148 D2 1N4148 C4 1uF C5 0.1uF L1 1uH C2 2 X 10TPB100ML, 100uF, 55mV C1 47uF Tantalum
C3 0.1uF
Vcc
Vc HDrv Q1 1/2 of IRF7301 L2 3.3V @ 4A C7 2 X 6TPC150M, 150uF, 40mV
C8 0.1uF
SS
U1 IRU3037
LDrv
D05022P-103, 10uH, 4.3A Q2 1/2 of IRF7301
C9 680pF R4 105K
Comp Fb Gnd R5 1K, 1%
R6 1.65K, 1%
Figure 18 - Typical application of IRU3037 in an on-board DC-DC converter using a single 5V supply.
14
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Rev. 2.3 03/06/02
IRU3037 / IRU3037A
TYPICAL APPLICATION
Dual Supply, 5V Bus and 12V Bias Input
5V L1 12V 1uH C2 10TPB100M, 100uF, 55m V , 1.5A rms C1 47uF 1.8V/1A IRU1206-18
Vcc V c HDrv SS
C4 0.1uF
C1 1uF
C3 47uF Q1 1/2 of IRF7752 L2 2.5V/2A
C7 0.1uF
U1 IRU3037
LDrv
CTX5-2P, 3.5uH @ 2.5A Q2 1/2 of IRF7752 R1
C6 6TPB150M, 150uF, 55m V
Cm op
C8 2200pF R2 14K
Fb
1K, 1%
Gnd
R3 1K, 1%
L2 CTX5-2P, 3.5uH @ 2.5A CTX10-5P, 5.7uH @ 2.5A
C6 6TPB150M, 150uF, 55m V (Qty 2) 6TPB150M, 150uF, 55m V (Qty 1)
C9 10TPB100M, 100uF, 55m V, 1.5A rms
C10 0.1uF
C11 1uF
Vcc
V c HDrv
Q3 1/2 of IRF7752
SS
C13 0.1uF
U1 IRU3037
LDrv
L3
3.3V/1.8A
CTX5-1P, 3.4uH @ 2A Q4 1/2 of IRF7752 R4
C12 6TPB150M, 150uF, 55m V
Cm op
C14 2200pF
Gnd
Fb
1.65K, 1% R5 14K R6 1K, 1%
Figure 19 - Typical application of IRU3037 or IRU3037A in an on-board DC-DC converter providing the Core, GTL+, and Clock supplies for the Pentium II microprocessor.
Rev. 2.3 03/06/02
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15
IRU3037 / IRU3037A
TYPICAL APPLICATION
1.8V to 7.5V / 0.5A Boost Converter
Vpwr (1.5V Min)
L1 1uH (CoilTronics UP2B-1R0)
Vcc
C5 1uF
C1 68uF 2X D1 1N5817
Vout (7.5V / 0.5A)
C9 47uF 2X
R1 20K R2 10K Q1 2N2222 R3 20K Q2 2N2222 C10 100pF C4 0.01uF R4 25K C5 0.1uF
Q3 IRF7402
SS Comp U1
Vc
HDrv
IRU3037
Fb Vcc LDrv Gnd
C8 1uF
Gnd
R5 1.00K 1% R6 5.00K 1%
Figure 20 - Typical application of IRU3037 as a boost converter.
16
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Rev. 2.3 03/06/02
IRU3037 / IRU3037A
DEMO-BOARD APPLICATION
5V or 12V to 3.3V @ 10A
L1
VIN 5V or 12V
1uH C1 33uF 16V D4 LL4148
C2A 47uF 16V
C2B 47uF 16V D2 LL4148
D1 LL4148
Gnd
C3 1uF
C19 1uF C6 Q1 IRF7457 0.1uF L2 3.3uH Q2 C7 470pF R4 4.7 V C9B 150uF 6.3V C9C 150uF 6.3V
Vcc
C8 1uF
Vc HDrv
SS
C5 0.1uF
U1 IRU3037
LDrv
C13 1uF
VOUT 3.3V @ 10A Gnd
Comp
C4 2200pF R3 20K
IRF7460 R6
Gnd
Fb
R5 1K
1.65K
Figure 21 - Demo-board application of IRU3037.
Application Parts List Ref Desig Q1 Q2 U1 D1, D2, D4 L1 L2 C1 C2A, C2B C9B, C9C C5, C6 C3 C4 C7 C8, C13, C19 R3 R4 R5 R6
Rev. 2.3 03/06/02
Description MOSFET MOSFET Controller Diode Inductor Inductor Capacitor, Tantalum Capacitor, Poscap Capacitor, Poscap Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Resistor Resistor Resistor Resistor
Value Qty Part# 20V, 7mV, 15A 1 IRF7457 20V, 10mV, 12A 1 IRF7460 Synchronous PWM 1 IRU3037 Fast Switching 3 LL4148 1mH, 10A 1 D03316P-102HC 3.3mH, 12A 1 D05022P-332HC 33mF, 16V 1 ECS-T1CD336R 47mF, 16V, 70mV 2 16TPB47M 150mF, 6.3V, 40mV 2 6TPC150M 0.1mF, Y5V, 25V 2 ECJ-2VF1E104Z 1mF, X7R, 25V 1 ECJ-3YB1E105K 2200pF, X7R, 50V 1 ECJ-2VB1H222K 470pF, X7R 1 ECJ-2VB2D471K 1uF, Y5V, 16V 3 ECJ-2VF1C105Z 20K, 5% 1 4.7V, 5% 1 1K, 1% 1 1.65K, 1% 1 www.irf.com
Manuf IR IR IR Coilcraft Coilcraft Panasonic Sanyo Sanyo Panasonic Panasonic Panasonic Panasonic Panasonic
17
IRU3037 / IRU3037A
DEMO-BOARD WAVEFORMS
IRU3037
Vin=5.0V, Vout=3.3V
100 Efficiency (%) 90 80
VIN
VOUT
70 0 1 2 3 4 5 6 7 8 9 10 11 Output Current (A)
Figure 22 - Efficiency for IRU3037 Evaluation Board.
Figure 23 - Start-up time @ IOUT=5A.
Vss
IRU3037 IRU3037
VOUT
IOUT = 5V
Figure 24 - Shutdown the output by pulling down the soft-start.
Figure 25 - 3.3V output voltage ripple @ IOUT=5A.
IRU3037
IRU3037
2A
4A
0A
0A
Figure 26 - Transient response @ IOUT = 0 to 2A.
Figure 27 - Transient response @ IOUT = 0 to 4A.
Rev. 2.3 03/06/02
18
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