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High Performance Quad Output Switching Regulator
POWER MANAGEMENT Description
The SC2453 is a high performance controller for multioutput converters that can be configured for a wide variety of applications. The SC2453 utilizes PWM synchronous buck topologies where efficiency is most important. It also provides dedicated programmable positive and negative linear regulators using external transistors. Each of the three positive outputs is adjustable down to 0.5V. The two PWM switchers are synchronized 180 out of phase reducing input ripple, allowing for fewer input capacitors. Power up sequencing prevents converter latchup. The SC2453 can be synchronized to other converters to prevent beat frequencies. The wide range programmable operating frequency allows users to optimize a converter design. The PWM switchers sense the voltage across the low-side MOSFETs on-resistance to efficiently provide adjustable current-limit, eliminating costly current-sense resistors. A POK signal is issued when soft-start is complete on both PWM switchers and their outputs are within 10% of the set point.
SC2453
Features
Two synchronized converters for low noise Power up sequencing to prevent latch-up Out of phase operation for low input ripple Over current protection Wide input range, 4.5 to 30V Programmable frequency up to 700KHz Two synchronous bucks for high efficiency at high current One dedicated programmable positive linear regulator and one dedicated programmable negative linear regulator Output voltage as low as 0.5V Small package TSSOP-28. Also offered in lead free, WEEE and RoHS compliant package
Applications
DSL applications with multiple input voltage requirements Mixed-Signal applications requiring positive and negative voltages Cable modem power management Base station power management
Typical Application Circuit
Vin: 4.5 - 30V
Q1 R1 C2
C1
C3 R3
C4 16 22 28 27
D1
D2 D3 C5 L1
A VCC
6 9
P VCC
V IN
B DI
C6
SS/SHDN OSC S Y NC POK
BS T2 GD2H PH2 GD2L FB2
17 18 19 20 13 12 26 25 24 23 2 3 C19 C20 C13 R8 C12
C7
Q2
Vout2: +3.3V
C8 R5
R4
7 8
To Vout2
C10
C9
Q3
(as low as 0.5V)
R6 C11
Vout4: +2.5V
R9 C14
Q4
R7
4
BD4
SC 2453
EO2 BS T1
5 R10 11
Q5
FB4
GD1H PH1
L2
Vout1: +5V
C15 R12
Q6
BD3 A GND P GND
GD1L FB1 ILI M2
(as low as 0.5V)
Vout3: -12V
R14
Q7
R11
C16 FB3 ILI M1 10
EO1
C17
C18
15
21
14
R15
R13
R16
1 R17
R18
To Vout2
Revision: July 20, 2005 1 www.semtech.com
SC2453
POWER MANAGEMENT Absolute Maximum Ratings
Exceeding the specifications below may result in permanent damage to the device, or device malfunction. Operation outside of the parameters specified in the Electrical Characteristics section is not implied.
Parameter BST1, BST2 to PGND VIN, BDI to AGND PVCC, AVCC to PGND PGND to AGND PH1 to BST1, PH2 to BST2 GD1H to PH1, GD2H to PH2 GD1L, GD2L to PGND ILIM1, ILIM2 to AGND OSC, SYNC, POK, SS to AGND BD3 to AVCC BD4 to AGND FB1, FB2, FB3, FB4, EO1, EO2 to AGND GD1H, GD1L, GD2H, DG2L Source or Sink Current Storage Temperature Range Junction Temperature Lead Temperature (Soldering) 10 Sec.
Maximum 35 30 7 0.3 -6 to 0.3 7 7 7 7 -25 to 0.3 7 -0.3 to 7 1 -60 to +150 -40 to +150 300
Units V V V V V V V V V V V V A C C C
Electrical Characteristics
Unless specified: VCC = 5V, fs = 600KHz, SS/SHDN = 5V, SYNC = 0V, TA = T = -35C to 105C
J
Parameter Pow er Supply Quiescent Current Operating Current AVCC PVC C Undervoltage Lockout Start Threshold UVLO Hysteresis PWM Comparator Delay to Output
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Test Conditions
Min
Typ
Max
Unit
SS/SHDN = 0V No load VIN > 5.5V VIN > 5.5V 4.5 4.5
3.8 9 5 5
4.5 12 5.5 5.5
mA mA V V
4.05
4.23 200
4.50
V mV
70
2
nS
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SC2453
POWER MANAGEMENT Electrical Characteristics (Cont.)
Unless specified: VCC = 5V, fs = 600KHz, SS/SHDN = 5V, SYNC = 0V, TA = T = -35C to 105C
J
Parameter Positive Linear Regulator Feedback Voltage Feedback Input Leakage Current Amplifier Transconductance Drive Sink Current Negative Linear Regulator Feedback Voltage Feedback Input Leakage Current Amplifier Trandconductance Driver Source Current PWM Error Amplifier Feedback Voltage Input Bias Current Input Offset Voltage Open Loop Gain Unity Gain Bandwidth Output Sink Current Output Source Current Slew Rate Oscillator Frequency Range Frequency CT Peak Voltage CT Valley Voltage SYNC Input High Pulse Width SYNC Rise/Fall Time SYNC Frequency Range SYNC High/Low Threshold
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Test Conditions
Min
Typ
Max
Unit
0.475
0.5 150
0.525
V nA
VFB4 = 0.5V, IBD4 = 0.5 to 5mA (sink) @ 1V output
-2 5
-10
mV mA
-25
0 150
25
mV nA
VFB3 = 0V, IBD3 = 0.5 to 5mA (source) @ -12V output
-5 5
-10
mV mA
0.49
0.5
0.51 200
V nA mV dB
3 90 3 2 2 1
6
MHz mA mA V/S
100 RT = 12.5K 540 600 3.0 0.1 100
700 660
KHz KHz V V nS
50 FOSC 1.5
3
nS kHz V
FOSC +10%
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SC2453
POWER MANAGEMENT Electrical Characteristics (Cont.)
Unless specified: VCC = 5V, fs = 600KHz, SS/SHDN = 5V, SYNC = 0V, TA = T = -35C to 105C
J
Parameter Duty Cycle PWM 1 & 2 Maximum Duty Cycle PWM 1 & 2 Maximum Duty Cycle PWM 1 & 2 Minimum Duty Cycle Current Limit CH1& 2 ILIM Set Voltage Soft Start/Shut Dow n Charge Current Discharge Current Disable Threshold Voltage Disable Low to Shut Down Output Gate Drive On-Resistance(H) Gate Drive On-Resistance(L) Rise Time(H) Fall Time(H) Rise Time(L) Fall Time(L) Pow er Good FB1 & FB2 Trip Level above FB Hysteresis FB1 & FB2 Trip Level below FB Hysteresis PWR OK Output Low Level PWR OK Output High Leakage
Test Conditions
Min
Typ
Max
Units
fs = 100kHz fs = 700kHz
90 71 2
% % %
1.8
2
2.2
V
5
10 1
15
A mA
0.45
0.5 50
0.55
V s
ISOURCE=15mA ISINK=15mA COUT = 1000pF COUT = 1000pF COUT = 1000pF COUT = 1000pF
2 2 15 25 15 15
5 5
nS nS nS nS
2
10 1
20
% %
-20
-10 1
-2
% %
Pullup resistor = 10K Pullup resistor = 10K
0.4
0.6 30
V A
Note: (1) This device is ESD sensitive. Use of standard ESD handling precautions is required.
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SC2453
POWER MANAGEMENT Pin Configuration
Top View
Ordering Information
Part Number SC2453ITSTR SC2453ITSTRT(2) Package(1) TSSOP-28 Temp. Range (TJ) -35C to +105C
Notes: (1) Only available in tape and reel packaging. A reel contains 2500 devices. (2) Lead free product. This product is fully WEEE and RoHS compliant.
(28 Pin TSSOP)
Marking Information
TOP
nnnn = Part Number (Example: 1406) yyww = Date Code (Example: 0012) xxxxx = Semtech Lot No. (Example: P94A01)
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SC2453
POWER MANAGEMENT Pin Descriptions
Pin # 1 2 Pin Name ILIM1 FB 1 Pin Function Adjustable current-limit pin set for the PWM switcher1. Connect a resistor from this pin to AGND. Voltage mode PWM switcher1 feedback input. Connect to a resistor-divider from output to FB1 to AGND to adjust the output voltage between 0.5V and 0.9 x VIN as maximum duty cycle allowed. The feedback set point is 0.5V. The PWM switcher1 loop compensation pin. Connect a compensation network to compensate the control loop. Type 3 compensation network is usually used. Open-drain output PNP transistor driver. Internally connected to the drain of an N-channel MOSFET. This pin connects to the base of an external PNP pass transistor to form a positive linear regulator. PNP transistor positive linear regulator feedback input. Connect to a resistor-divider between the positive linear regulator and AGND to adjust the output voltage. The feedback set point is 0.5V. The pin provides a soft-start function for the PWM switchers and positive linear regulator. When the pin reaches 0.5V, the regulated 10uA pull-up current source charges the capacitor connected from this pin to AGND. The error amplifier reference voltage of switcher1, switcher2 and positive linear regulators ramps from 0.7V to 2V respectively following the sequence. If the pin is pulled down below 0.5V, the SC2453 is disabled. The pin can be used to synchronize two or more controllers. This pin requires a 10K resistor to AGND when it is not used. Open-drain power-good output. POK is low when the output voltage is more than 10% below or above the regulation point. POK is high impedance when the output is in regulation. Connect a resistor between the pin and PVCC. Oscillator frequency adjustable input. Connect a resistor between the pin and AGND to set the PWM frequency. NPN transistor negative linear regulator feedback input. Connect to a resistor-divider between a positive reference voltage and the negative output to adjust the output voltage. The feedback set point is 0V. Open-drain output NPN transistor driver. Internally connected to the drain of a P-channel MOSFET. This pin connects to the base of an external NPN pass transistor to form a negative linear regulator. The PWM switcher2 loop compensation pin. Connect a compensation network to compensate the control loop. Type 3 compensation network is usually used. Voltage mode PWM switcher2 feedback input. Connect to a resistor-divider from output to FB2 to AGND to adjust the output voltage between 0.5V and 0.9 x V IN as maximum duty cycle allowed. The feedback set point is 0.5V. Adjustable current-limit pin set for the PWM switcher2. Connect a resistor from this pin to AGND. This is small signal ground and must be routed separately from the high current ground PGND. All voltage levels are measured with respect to this pin. A ceramic capacitor should be connected right to this pin for noise decoupling.
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3 4
EO1 BD 4
5
FB 4
6
SS/SHDN
7 8
SYNC POK
9 10
OSC FB 3
11
BD 3
12 13
EO2 FB 2
14 15
ILIM2 AGND
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SC2453
POWER MANAGEMENT Pin Descriptions (Cont.)
Pin # 16 Pin Name AVCC Pin Function This pin is connected to the output of the external linear regulator to supply the IC. The IC may be powered directly from a single 5V (10%) supply at this pin. The pin must be always decoupled to PGND with a minimum of 1uF ceramic capacitor, place very close to the pin. The pin powers the upper MOSFET driver of PWM switcher2. Connect this pin to the junction of the bootstrap capacitor and cathode of the bootstrap diode. The anode of the bootstrap diode is connected to the PVCC. Switcher2 upper MOSFET gate driver output. It swings between BST2 and PH2. The pin is connected to the junction of the upper MOSFET's source, output inductor and lower MOSFET's drain. Switcher2 lower MOSFET gate driver output. It swings between PVCC and PGND. The pin provides the power ground for the lower gate drivers for both PWM switchers. It should be connected to the sources of the lower MOSFETs and the negative terminal of the input capacitors. This pin is connected to the output of the external linear regulator to supply the internal lower gate drivers for both PWM switchers. It may be powered directly from a single 5V (10%) supply at this pin. The pin must be always decoupled to PGND with a minimum of 10uF ceramic capacitor, place very close to the pin. Switcher1 lower MOSFET gate driver output. It swings between PVCC and PGND. The pin is connected to the junction of the upper MOSFETs source, output inductor and lower MOSFETs drain. Switcher1 upper MOSFET gate driver output. It swings between BST1 and PH1. The pin powers the upper MOSFET driver of PWM switcher1. Connect this pin to the junction of the bootstrap capacitor and cathode of the bootstrap diode. The anode of the bootstrap diode is connected to the PVCC. The pin is internally connected to the pull-up resistor of an N-channel MOSFET to regulate the bias for the IC. The external bias pass PNP transistor driver. Internally connected to the drain of an N-channel MOSFET.
17
BST2
18 19 20 21
GD2H PH2 GD2L PGND
22
PVC C
23 24 25 26
GD1L PH1 GD1H BST1
27 28
VIN BD I
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SC2453
POWER MANAGEMENT Block Diagram
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SC2453
POWER MANAGEMENT Applications Information
The SC2453 is designed to control and drive two N-Channel MOSFET PWM synchronous buck switchers and two positive linear regulators. The two PWM switchers are synchronized 180 out of phase for low input ripple and noise. The switching frequency is programmable to optimize design. The SC2453 PWM switchers feature lossless current sensing and programmable over current limit. The two positive linear regulators output voltages are adjustable. Power Supplies Supplies VIN, PVCC and AVCC from the input source are used to power the SC2453. An external PNP transistor linear regulator supplies AVCC and PVCC. The AVCC supply provides the bias for the oscillator, the switchers, the linear regulator controllers and the POK circuitry. PVCC is used to drive the low side MOSFET gate. In low shutdown current mode, the PNP transistor is turned off, disabling AVCC and PVCC. Soft-start, Sequencing and Disabling A 10A current source pulls up on the SS/SHDN pin. When the SS/SHDN pin reaches 0.5V, the first switcher is activated and the reference input of the error amplifier is ramped up with the soft-start voltage. When the SS/ SHDN pin reaches 2V, the SS/SHDN pin is pulled down to approximately 0.7V and the second switcher begins to soft-start in an identical fashion to the first switcher. When the SS/SHDN pin reaches 2V for the second time, the SS/SHDN pin is pulled down to approximately 0.7V again, and then the positive linear regulators ramp up with the SS/SHDN pin voltage. The SS/SHDN pin is eventually pulled up to the supply AVCC. The soft-start time is controlled by the value of the capacitor connected to the SS/SHDN pin. If the SS/SHDN pin is pulled down below 0.5V, the SC2453 is disabled. If the SS/SHDN pin is pulled down below 0.34V, the bias PNP transistor for SC2453 is disabled and the supply current is only 100uA. The power-ok circuitry monitors the FB inputs of the error amplifiers of the switchers. If the voltage on these inputs goes above 0.55V or below 0.45V then the POK pin is pulled low. The POK pin is held low until the end of the start-up sequence.
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Frequency Setting and Synchronization The internal oscillator free-running frequency of the SC2453 is set by an external resistor using the following formula:
R freq = 7.9 x 10 9 fs - 12 x 10 3
When it is synchronized externally, the applied clock frequency should be equal or greater than the free-running frequency. Setting Current Limit SC2453 monitors the voltage drop in the lower MOSFETs Rdson voltage to sense an over current condition. This method of current sensing minimizes any unnecessary losses due to external sense resistance. The SC2453 utilizes an internal current source and an external resistor connected from the ILIM pins to the AGND pin to program a current limit level. This limit is programmable by choosing the resistor relative to the level required. The value of the resistor can be selected by the following formula:
Ri lim = 2000 /(IIim * Rdson)
Rilim should be between 10K and 100K. An internal comparator with a reference from the level set by the external resistor monitors the voltage drop across the lower MOSFET. Once the Vdson of the MOSFET exceeds this level, the low side gate is turned on and the upper MOSFET is turned off in the next switching cycle. Gate Drives The low side gate driver is supplied from PVCC and provides a peak source/sink current of 1A. The high side gate drive is also capable of sourcing and sinking peak currents of 1A. The high side MOSFET gate drive can be provided by an external 12V supply that is connected from BST to GND. The actual gate to source voltage of the upper MOSFET will approximately equal 7V (12V-VCC). If the external 12V supply is not available, a classical bootstrap technique can be implemented from the PVCC supply. A bootstrap capacitor is connected from BST to Phase while PVCC is connected through a diode (Schottky or other fast low VF diode) to the BST. This will provide a gate to source voltage approximately equal to the VCCwww.semtech.com
SC2453
POWER MANAGEMENT Applications Information (Cont.)
Vdiode drop. Shoot through control circuitry provides a 30ns dead time to ensure both the upper and lower MOSFET will not turn on simultaneously and cause a shoot through condition. Error Amplifier and PWM Controller In closed loop operation, the internal oscillator ramp ranges from 0.75V to 3.8V. The error amplifier output ranges determines duty-ratio of a converter. The synchronous continuous-conduction mode of operation allows the SC2453 to regulate the output irrespective of the direction of the load current. The SC2453 uses voltage-mode control for good noise immunity and ease of compensation. The low-side MOSFET of each channel is turned off at the falling-edge of the phase timing clock. After a brief non-overlapping time interval of 30ns, the high-side MOSFET is turned on. The phase inductor current ramps up. When the internal ramp reaches the threshold determined by the error amplifier output, the high-side MOSFET is turned off. As long as phase voltage collapses below 1.5V, the low-side MOSFET is turned on. Buck Converter Buck converter design includes the following specifications: Input voltage range: Vin [ Vin,min , Vin,max ] Input voltage ripple (peak-to-peak): Vin Output voltage: Vo Output voltage accuracy: Output voltage ripple (peak-to-peak): Vo Nominal output (load) current: Io Maximum output current limit: Io,max Output (load) current transient slew rate: dIo (A/s) Circuit efficiency: Selection criteria and design procedures for the following are described. 1) output inductor (L) type and value, 2) output capacitor (Co) type and value, 3) input capacitor (Cin) type and value, 4) power MOSFET's, 5) current sensing and limiting circuit, 6) voltage sensing circuit, 7) loop compensation network.
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Operating Frequency (f s ) The switching frequency in the SC2453 is userprogrammable. The advantages of using constant frequency operation are simple passive component selection and ease of feedback compensation. Before setting the operating frequency, the following trade-offs should be considered. 1) 2) 3) 4) 5) Passive component size Circuitry efficiency EMI condition Minimum switch on time and Maximum duty ratio
For a given output power, the sizes of the passive components are inversely proportional to the switching frequency, whereas MOSFET's/Diodes switching losses are proportional to the operating frequency. Other issues such as heat dissipation, packaging and the cost issues are also to be considered. The frequency bands for signal transmission should be avoided because of EM interference. Minimum Switch On Time Consideration In the SC2453, the falling edge of the clock turns on the top MOSFET. The inductor current and the sensed voltage ramp up. After the internal ramp voltage crosses a threshold determined by the error amplifier output, the top MOSFET is turned off. The propagation delay time from the turn-on of the controlling FET to its turn-off is the minimum switch on time. The SC2453 has a minimum on time of about 50ns at room temperature. This is the shortest on interval of the controlling FET. The controller either does not turn on the top MOSFET at all or turns it on for at least 50ns. For a synchronous step-down converter, the operating duty cycle is VO/VIN. So the required on time for the top MOSFET is VO/(VINfs). If the frequency is set such that the required pulse width is less than 50ns, then the converter will start skipping cycles. Due to minimum on time limitation, simultaneously operating at very high switching frequency and very short duty cycle is not practical. If the voltage conversion ratio VO/VIN and hence the required duty cycle is higher, the switching frequency can be increased to reduce the size of passive components.
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SC2453
POWER MANAGEMENT Applications Information (Cont.)
There will not be enough modulation headroom if the on time is simply made equal to the minimum on time of the SC2453. For ease of control, we recommend the required pulse width to be at least 1.5 times the minimum on time. I nductor (L) and Ripple Current Both step-down controllers in the SC2453 operate in synchronous continuous-conduction mode (CCM) regardless of the output load. The output inductor selection/design is based on the output DC and transient requirements. Both output current and voltage ripples are reduced with larger inductors but it takes longer to change the inductor current during load transients. Conversely smaller inductors results in lower DC copper losses but the AC core losses (flux swing) and the winding AC resistance losses are higher. A compromise is to choose the inductance such that peak-to-peak inductor ripple-current is 20% to 30% of the rated output load current. Assuming that the inductor current ripple (peak-to-peak) value is *Io, the inductance value will then be:
L= Vo (1 - D ) Io fs
c) Current rating: The saturation current of the inductor should be at least 1.5 times of the peak inductor current under all conditions. Output Capacitor (C o ) and V out Ripple The output capacitor provides output current filtering in steady state and serves as a reservoir during load transient. The output capacitor can be modeled as an ideal capacitor in series with its parasitic ESR (Resr) and ESL (Lesl) (Figure 1).
Co
Lesl
Resr
Figure 1. An equivalent circuit of Co. If the current through the branch is ib(t), the voltage across the terminals will then be:
v o ( t ) = Vo + 1 di ( t ) ib ( t )dt + L esl b + R esr ib ( t ) Co 0 dt
The peak current in the inductor becomes (1+/2)*Io and the RMS current is:
IL,rms = Io 1 + 2 12
t
This basic equation illustrates the effect of ESR, ESL and Co on the output voltage. The first term is the DC voltage across Co at time t=0. The second term is the voltage variation caused by the charge balance between the load and the converter output. The third term is voltage ripple due to ESL and the fourth term is the voltage ripple due to ESR. The total output voltage ripple is then a vector sum of the last three terms. Since the inductor current is a triangular waveform with peak-to-peak value *Io, the ripple-voltage caused by inductor current ripples is:
v C Io 8 C o fs Io D
The followings are to be considered when choosing inductors. a) Inductor core material: For high efficiency applications above 350KHz, ferrite, Kool-Mu and polypermalloy materials should be used. Low-cost powdered iron cores can be used for cost sensitive-applications below 350KHz but with attendant higher core losses. b) Select inductance value: Sometimes the calculated inductance value is not available off-the-shelf. The designer can choose the adjacent (larger) standard inductance value. The inductance varies with temperature and DC current. It is a good engineering practice to re-evaluate the resultant current ripple at the rated DC output current.
the ripple-voltage due to ESL is:
v ESL = L esl fs
and the ESR ripple-voltage is:
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SC2453
POWER MANAGEMENT
v ESR = R esr Io
Aluminum capacitors (e.g. electrolytic, solid OS-CON, POSCAP, tantalum) have high capacitances and low ESL's. The ESR has the dominant effect on the output ripple voltage. It is therefore very important to minimize the ESR. When determining the ESR value, both the steady state ripple-voltage and the dynamic load transient need to be considered. To keep the steady state output ripple-voltage < Vo, the ESR should satisfy:
R esr 1 < Vo Io
To limit the dynamic output voltage overshoot/ undershoot within (say 3%) of the steady state output voltage) from no load to full load, the ESR value should satisfy:
R esr 2 < Vo Io
capacitor value. Only when the capacitor value is set properly, the effect of ceramic capacitor low ESR starts to be significant. For example, if a 10F, 4m ceramic capacitor is connected in parallel with 2x1500F, 90m electrolytic capacitors, the ripple current in the ceramic capacitor is only about 42% of the current in the electrolytic capacitors at the ripple frequency. If a 100F, 2m ceramic capacitor is used, the ripple current in the ceramic capacitor will be about 4.2 times of that in the electrolytic capacitors. When two 100F, 2m ceramic capacitors are used, the current ratio increases to 8.3. In this case most of the ripple current flows in the ceramic decoupling capacitor. The ESR of the ceramic capacitors will then determine the output ripple-voltage. Remark 2: The total equivalent capacitance of the filter bank is not simply the sum of all the paralleled capacitors. The total equivalent ESR is not simply the parallel combination of all the individual ESR's either. Instead they should be calculated using the following formulae.
C eq () := (R1a + R1b )2 2C1a C1b + (C1a + C1b )2 (R1a C1a + R1b C1b )2 C1a C1b + (C1a + C1b ) R1aR1b (R1a + R1b )2C1a C1b + (R1b C1b + R1a C1a ) (R1a + R1b )2 2 C1a C1b + (C1a + C1b )2
2 2 2 2 2 2 2 2 2 2
Then, the required ESR value of the output capacitors should be: Resr = min{Resr1,Resr2 } The voltage rating of aluminum capacitors should be at least 1.5Vo. The RMS current ripple rating should also be greater than:
Io 23
R eq () :=
Usually it is necessary to have several capacitors of the same type in parallel to satisfy the ESR requirement. The voltage ripple cause by the capacitor charge/ discharge should be an order of magnitude smaller than the voltage ripple caused by the ESR. To guarantee this, the capacitance should satisfy:
Co > 10 2 fsR esr
where R 1a and C 1a are the ESR and capacitance of electrolytic capacitors, and R1b and C1b are the ESR and capacitance of the ceramic capacitors respectively. (Figure 2)
C1a
C1b
Ceq
In many applications, several low ESR ceramic capacitors are added in parallel with the aluminum capacitors in order to further reduce ESR and improve high frequency decoupling. Because the values of capacitance and ESR are usually different in ceramic and aluminum capacitors, the following remarks are made to clarify some practical issues. Remark 1: High frequency ceramic capacitors may not carry most of the ripple current. It also depends on the
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R1a
R1b
Req
Figure 2. Equivalent RC branch. Req and Ceq are both functions of frequency. For rigorous design, the equivalent ESR should be evaluated at the ripple frequency for voltage ripple calculation when both ceramic and electrolytic capacitors are used. If R1a = R1b
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SC2453
POWER MANAGEMENT
= R1 and C1a = C1b = C1, then Req and Ceq will be frequencyindependent and Req = 1/2 R1 and Ceq = 2C1 Input Capacitor (C in) The input supply to the converter usually comes from a pre-regulator. Since the input supply is not ideal, input capacitors are needed to filter the current pulses at the switching frequency. A simple buck converter is shown in Figure 3. inductances around the pulse current loop. The RMS value of the capacitor current is approximately
ICin = Io D[(1 + 2 D D )(1 - )2 + 2 (1 - D) ] 12
The power dissipated in the input capacitors is then: PCin = ICin2Resr For reliable operation, the maximum power dissipation in the capacitors should not result in more than 10oC of temperature rise. Many manufacturers specify the maximum allowable ripple current (ARMS) rating of the capacitor at a given ripple frequency and ambient temperature. The input capacitance should be high enough to handle the ripple current. For higher power applications, multiple capacitors are placed in parallel to increase the ripple current handling capability. Sometimes meeting tight input voltage ripple specifications may require the use of larger input capacitance. At full load, the peak-to-peak input voltage ripple due to the ESR is:
v ESR = R esr (1 + )Io 2
Figure 3. A simple model for the converter input In Figure 3 the DC input voltage source has an internal impedance Rin and the input capacitor Cin has an ESR of Resr. MOSFET and input capacitor current waveforms, ESR voltage ripple and input voltage ripple are shown in Figure 4.
The peak-to-peak input voltage ripple due to the capacitor is:
v C DI o C in fs
From these two expressions, CIN can be found to meet the input voltage ripple specification. In a multi-phase converter, channel interleaving can be used to reduce ripple. The two step-down channels of the SC2453 operate at 180 degrees from each other. If both stepdown channels in the SC2453 are connected in parallel, both the input and the output RMS currents will be reduced. Ripple cancellation effect of interleaving allows the use of smaller input capacitors. When converter outputs are connected in parallel and interleaved, smaller inductors and capacitors can be used for each channel. The total output ripple-voltage remains unchanged. Smaller inductors speeds up output load transient. When two channels with a common input are interleaved, the total DC input current is simply the sum of the individual DC input currents. The combined input current waveform depends on duty ratio and the output current
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Figure 4. Typical waveforms at converter input. It can be seen that the current in the input capacitor pulses with high di/dt. Capacitors with low ESL should be used. It is also important to place the input capacitor close to the MOSFET's on the PC board to reduce trace
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SC2453
POWER MANAGEMENT
waveform. Assuming that the output current ripple is small, the following formula can be used to estimate the RMS value of the ripple current in the input capacitor.
Gate Charge (nC) 50 40 Cg( 100 , Rds) Cg( 200 , Rds) Cg( 500 , Rds) 20
Let the duty ratio and output current of Channel 1 and Channel 2 be D1, D2 and Io1, Io2, respectively. If D1<0.5 and D2<0.5, then:
ICin D I
2 1 o1
1
0
+ D 2Io 2
2
0 1
5
10
15
20 20
If D1>0.5 and (D1-0.5) < D2<0.5, then:
ICin 0.5Io1 + (D1 - 0.5 )(Io1 + Io 2 )2 + (D 2 - D1 + 0.5 )Io 2
2 2
FOM:100*10^{-12} FOM:200*10^{-12} FOM:500*10^{-12}
Rds On-resistance (mOhm)
Figure 5. Figure of Merit curves. MOSFET selection also depends on applications. In many applications, either switching loss or conduction loss dominates for a particular MOSFET. For synchronous buck converters with high input to output voltage ratios, the top MOSFET is hard switched but conducts with very low duty cycle. The bottom switch conducts at high duty cycle but switches at near zero voltage. For such applications, MOSFET's with low Cg are used for the top switch and MOSFET's with low Rds(on) are used for the bottom switch. MOSFET power dissipation consists of a) conduction loss due to the channel resistance Rds(on), b) switching loss due to the switch rise time tr and fall time tf, and c) the gate loss due to the gate resistance RG. Top Switch: The RMS value of the top switch current is calculated as:
IQ1,rms = Io D(1 +
2 12
If D1>0.5 and D2 < (D1-0.5) < 0.5, then:
ICin 0.5Io1 + D 2 (Io1 + Io 2 )2 + (D1 - D 2 - 0.5 )Io 2
2 2
If D1>0.5 and D2 > 0.5, then:
ICin (D1 + D 2 - 1)(Io1 + Io 2 )2 + (1 - D 2 )Io1 + (1 - D1 )Io 2
2 2
Choosing Power MOSFET's Main considerations in selecting the MOSFET's are power dissipation, cost and packaging. Switching losses and conduction losses of the MOSFET's are directly related to the total gate charge (Cg) and channel on-resistance (Rds(on)). In order to judge the performance of MOSFET's, the product of the total gate charge and on-resistance is used as a figure of merit (FOM). Transistors with the same FOM follow the same curve in Figure 5. The closer the curve is to the origin, the lower is the FOM. This means lower switching loss or lower conduction loss or both. It may be difficult to find MOSFET's with both low Cg and low Rds(on. Usually a trade-off between Rds(on and Cg has to be made.
)
The conduction losses are then: Ptc = IQ1,rms2 Rds(on) Rds(on) varies with temperature and gate-source voltage. Curves showing R ds(on) variations can be found in manufacturers' data sheet. From the Si4860 datasheet, Rds(on) is less than 8mW when Vgs is greater than 10V. However R ds(on) increases by 50% as the junction temperature increases from 25oC to 110oC. The switching losses can be estimated using the simple formula
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SC2453
POWER MANAGEMENT
Pts =
1 2 ( t r + t f )(1 + 2 )Io Vin fs
where tr is the rise time and tf is the fall time of the switching process. Different manufactures have different definitions and test conditions for t and t . To clarify r f these, we sketch the typical MOSFET switching characteristics under clamped inductive mode in Figure 6.
specified in the datasheet. The power dissipated within the MOSFET package is:
Ptg = Rg R gt Q g Vcc fs
The total power loss of the top switch is then: Pt = Ptc+Pts+Ptg If the input supply of the power converter varies over a wide range, then it will be necessary to weigh the relative importance of conduction and switching losses. This is because conduction losses are inversely proportional to the input voltage. Switching loss however increases with the input voltage. The total power loss of MOSFET should be calculated and compared for high-line and low-line cases. The worst case is then used for thermal design. Bottom Switch: The RMS current in bottom switch can be shown to be:
Figure 6. MOSFET switching characteristics Where, Qgs1 is the gate charge needed to bring the gate-to-source voltage Vgs to the threshold voltage Vgs_th, Qgs2 is the additional gate charge required for the switch current to reach its full-scale value Ids and . Qgd is the charge needed to charge gate-to-drain (Miller) capacitance when Vds is falling. Switching losses occur during the time interval [t1, t3]. Defining tr = t3-t1 and tr can be approximated as:
tr = (Q gs 2 + Q gd )R gt Vcc - Vgsp
IQ 2,rms = Io (1 - D )(1 +
2 12
)
The conduction losses are then: Pbc=IQ2,rms2 Rds(on) where Rds(on) is the channel resistance of bottom MOSFET. If the input voltage to output voltage ratio is high (e.g. Vin=12V, Vo=1.5V), the duty ratio D will be small. Since the bottom switch conducts with duty ratio (1-D), the corresponding conduction losses can be quite high. Due to non-overlapping conduction between the top and the bottom MOSFET's, the internal body diode or the external Schottky diode across the drain and source terminals always conducts prior to the turn on of the bottom MOSFET. The bottom MOSFET switches on with only a diode voltage between its drain and source terminals. The switching loss is:
1 Pbs = 2 ( t r + t f )(1 + 2 )Io Vd fs
where Rgt is the total resistance from the driver supply rail to the gate of the MOSFET. It includes the gate driver internal impedance Rgi, external resistance Rge and the gate resistance Rg within the MOSFET i.e.: Rgt = Rgi+Rge+Rg Vgsp is the Miller plateau voltage shown in Figure 11. Similarly an approximate expression for tf is:
is negligible due to near zero-voltage switching. The gate losses are estimated as :
Pbg = Rg R gt Q g Vcc fs
tf =
(Q gs 2 + Q gd )R gt Vgsp
The total bottom switch losses are then: Pb=Pbc+Pbs+Pbg
Only a portion of the total losses Pg = QgVccfs is dissipated in the MOSFET package. Here Qg is the total gate charge
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POWER MANAGEMENT
Once the power losses Ploss for the top (P t) and bottom (Pb) MOSFET's are known, thermal and package design at component and system level should be done to verify that the maximum die junction temperature (Tj,max, usually 125oC) is not exceeded under the worstcase condition. The equivalent thermal impedance from junction to ambient (ja) should satisfy:
ja Tj,max - Ta,max Ploss
Vo (V) (1- h)/h 0.6 0.2 0.9 0.8 806 1K 1.2 1.4 1.4K 1K 1.5 2 2K 1K 1.8 2.6 2.5 4 3.3 5.6
Ro1 (Ohm) 200 Ro2 (Ohm) 1K
2.61K 4.02K 5.62K 1K 1K 1K
ja depends on the die to substrate bonding, packaging material, the thermal contact surface, thermal compound property, the available effective heat sink area and the air flow condition (free or forced convection). Actual temperature measurement of the prototype should be carried out to verify the thermal design. Setting the Output Voltage The non-inverting input of the channel-one error amplifier is internally tied the 0.5V voltage reference output. A simple voltage divider (Ro1 at top and Ro2 at bottom) sets the converter output voltage. The voltage feedback gain h=0.5/Vo is related to the divider resistors value as:
Ro2 h = R o1 1- h
and its multiples fall into the standard resistor value chart (1%, 5% or so), it is possible to use standard value resistors to exactly set up the required output voltage value. The input bias current of the error amplifier also causes an error in setting the output voltage. The maximum inverting input bias currents of error amplifiers 1 or 2 is 200nA. Since the non-inverting input is biased to 0.5V, the percentage error in the second output voltage will be 100%*(0.2uA) R *R /[0.5 * (R +R ) ]. 01 o2 o1 o2 Valley Current Sensing for Current-Limit The valley current sensing for current limiting is a unique scheme which could sense the voltage across the bottom switch MOSFET when it is on. The scheme is robust with good noise immunity due to reference to ground. The current sensing point is at a delay time tdv before the beginning of a switching cycle. Therefore, the actual valley current is:
IV = - Vo RDS (ON ) _ B + RL (1 - e
- t dv 2
Once either R o1 or R o2 is chosen, the other can be calculated for the desired output voltage Vo. Since the number of standard resistance values is limited, the calculated resistance may not be available as a standard value resistor. As a result, there will be a set error in the converter output voltage. This non-random error is caused by the feedback voltage divider ratio. It cannot be corrected by the feedback loop. The following table lists a few standard resistor combinations for realizing some commonly used output voltages. Only the voltages in boldface can be precisely set with standard 1% resistors. From this table, one may also observe that when the value:
1 - h Vo - 0.5 = h 0.5
) + I VS e
-
t dv 2
where, IVS is the preset valley current limiting threshold. If a sensed current exceeds the threshold, the top switch will keep off in the next cycle until the current goes back below the threshold. In steady state, since the output voltage is out of regulation in over current condition, the control loop will try to make maximum duty cycle for the top switch as it is on, which is usually greater than 80%. Therefore, as the current falls back below the threshold, it is on in the next almost full cycle. The peak current is not controlled and only depends upon circuit parameters and operating condition in this cycle. The peak current IP is:
IP =
- - Vin - Vo (1 - e 1 ) + I V e 1 R DS (ON ) _ T + RL T T
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SC2453
POWER MANAGEMENT
Where, T is switching cycle period. Because the inductor current ramps up according to Vin -Vo, and down with Vo, a falling output voltage in over current condition moves the inductor "ramp up" current up faster and the peak current IP larger but the "ramp down" current slower. As long as the current is still larger than the current limiting threshold, the top switch is held off. The switching frequency is folded back lower but average current is relatively still high. An example is shown in Typical Characteristics section with input voltage 12V and output voltage 5V with output current 3.5A. The preset valley current limit is 5A, which has to be greater than full load current 3.5. When the output is shorted, the inductor peak current and the load current, both are a function of the input voltage and the output inductance, would be higher than the current limit. Therefore, proper thermal management and inductor saturation current rating have to be considered at the worst conditions. Loop Compensation SC2453 is a voltage mode buck controller with high gain bandwidth error amplifier utilizing external network compensation to regulate output voltage. For a DC/DC converter, it is usually required that the converter has a loop gain of a high crossover frequency for fast load response, high DC and low frequency gain for low steady state error, and enough phase margin for its operating stability. Often one can not have all these properties at the same time. The purpose of the loop compensation is to arrange the poles and zeros of the compensation network to meet the requirements for a specific application. The power stage of a buck converter control-to-output transfer function is as shown below:
G VD ( s ) = VIN 1 + sR CC L 1 + s + s 2LC R
C - Output capacitance RC - Output capacitor ESR VIN - Input voltage The transfer function of the error amplifier with external compensation network is as follows:
GCOMP (s ) = I s (1 + s )(1 + Z1 s (1 + )(1 + P1 s ) Z 2 s ) P 2
where,
Z1 = I = 1 1 , Z 2 = R 2 C1 (R1 + R 3 )C 2 1 C1C 3 R2 C1 + C 3
1 1 , P1 = , P 2 = R 1 ( C1 + C 3 ) R3C2
C3 C2 R1 + Vref R3 R2 C1
Figure 7. Voltage Mode Buck Converter Compensation Network With the compensation, the converter total loop gain is as follows:
s 1+ 1 s s 1 I VI 1 + 1+ RC C 4 Z1 Z 2 VM T(s) = GPWM GCOMP (s) G VD (s) = L s s s 1+ 1+ 1 + s 1 + s 2L1C P1 P 2 R
Where: GPWM = PWM gain VM = 3.0V, ramp peak to valley voltage of SC2453 The design guidelines are as following: 1. Set the loop gain crossover frequency C less than1/ 5th the switching frequency. 2. Place an integrator in the origin to increase DC and low frequency gains.
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where,
O = 1 LC
L - Output inductance
2005 Semtech Corp.
SC2453
POWER MANAGEMENT
3. Select Z1 and Z2 such that they are placed near O to dampen peaking; the loop gain has -20dB rate to go across the 0dB line for obtaining a wide bandwidth. 4. Cancel ESR with compensation pole P1 (P1 = ESR ). 5. Place a high frequency compensation pole P2 at the half switching frequency to get the maximum attenuation of the switching ripple and the high frequency noise with the adequate phase lag at C. The compensated loop gain will be as given in Figure 8:
T (s) z1 G vd 0d B P o w e r st ge a G V D (s) E S R -4 0d B / c de o z2 c p 1 p 2 -2 0d B / c de Lo op ga i T (s) n
The maximum voltage to drive an NPN transistor is AVCC minus the voltage drop across the internal P-MOSFET which is the product of On-Resistance and sourcing current. The maximum driving voltage with 5mA sourcing current is minimum AVCC (4.5V) minus 5mA times maximum On-Resistance 140, i.e. 3.8V. Layout Guidelines In order to achieve optimal electrical, thermal and noise performance for high frequency converters, attention must be paid to the PCB layouts. The goal of layout optimization is to place components properly and identify the high di/dt loops to minimize them. The following guideline should be used to ensure proper functions of the converters. 1. A ground plane is recommended to minimize noises and copper losses, and maximize heat dissipation. 2. Start the PCB layout by placing the power components first. Arrange the power circuit to achieve a clean power flow route. Put all the connections on one side of the PCB with wide copper filled areas if possible. 3. The PVCC and AVCC bypass capacitors should be placed next to the PVCC, AVCC and PGND, AGND pins respectively. 4. Separate the power ground from the signal ground. In SC2453, the power ground PGND should be tied to the source terminal of lower MOSFETs. The signal ground AGND should be tied to the negative terminal of the output capacitor. 5. The trace connecting the feedback resistors to the output should be short, direct and far away from the noise sources such as switching node and switching components. Minimize the traces between DRXH/ DRXL and the gates of the MOSFETs to reduce their impedance to drive the MOSFETs. 7. Minimize the loop including input capacitors, top/bottom MOSFETs. This loop passes high di/dt current. Make sure the trace width is wide enough to reduce copper losses in this loop. 8. Maximize the trace width of the loop connecting the inductor, bottom MOSFET and the output capacitors. 9. Connect the ground of the feedback divider and the compensation components directly to the GND pin of the SC2453 by using a separate ground trace. Then connect this pin to the ground of the output capacitor as close as possible.
Figure 8. Asymptotic diagram of buck power stage and its compensated loop gain. Positive and Negative LDO Controllers The SC2453 provides positive and negative adjustable linear regulator controllers. The positive linear regulator uses a PNP transistor to regulate output voltage. This is set by a voltage divider connected from the output to FB to AGND. Referring to the front page Application Circuit, select R10 in the 5K to 20K range. Calculate R9 with the following equation:
V R9 = R10 OUT - 1 0.5
The negative linear regulator uses a NPN transistor to regulate output voltage. This is set by a voltage divider connected from the output to FB to a positive reference. Referring to the front page Application Circuit, select R18 in the 5K to 20K range. Calculate R14 with the following equation:
V 14 = 18 OUT V REF
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16
22
28
27
BDI
VIN
AVCC
PVCC
AGND
PGND
ILIM2
1
15
21
GND
R22 51K C25 4.7nF R21 20K R23 51K R24 10K TP13
100pF
R20 36K 14 680pF
10uF
C53 10uF
ILIM1
2005 Semtech Corp.
12V GND
TP1 C5 10uF D1 SL04 D3 B180 C7 10uF 4 TP4 BST2 GD2H PH2 GD2L FB2 4 BD4 SC2453 BST1 TP8 5 FB4 PH1 11 BD3 FB1 10 FB3 C24 EO1 3 R19 2.0K 2 0 GD1L 23 R14 TP11 Q6 SUD50N03-10CP 24 TP12 GD1H R13 10K C18 Open R15 C21 Open 0 R17 1K C23 TP10 25 C15 390pF C16 0.1uF R12 2 Q5 SUD50N03-10CP L2 10uH 26 5.6nF EO2 12 R9 20K C14 R10 2.05K TP9 C19 3.3nF R16 18K R18 150 13 0 20 R6 TP6 19 Q3 SUD50N03-10CP 2 R5 11.5K R7 180 C10 6.8nF TP5 18 1 3 L1 STQ127_1042 R4 17 2 TP3 C9 0.1uF Q2 SUD50N03-10CP D2 SL04 C32 22uF R1 10K R2 10 Q1 FZT749 C50 1uF R50 20K R52 0 C1 10uF/16V C2 10uF/16V C3 10uF/16V C4 10uF/16V C27 Open C28 Open C29 Open C33 Open
POK C6 1uF
SYNC C8 0.1uF 6 SS/SD OSC SYNC POK 9 R3 30K 7 8 C31 2.2nF TP7 fs = 2 5 0 K H z
+3.3V/2A 3.3V
C11 680uF/4V C30 22uF
POWER MANAGEMENT Evaluation Board Schematic
SS/SD R51 10K
GND
+3.3V/2A
C12 Open
C13 10uF
R8
GND
Q4 FZT749
1K
2.5V/1A
Q7 FZT649
C17 10uF
R11 40.2K
5V/2
C20 C51 470uF/6.3V 22uF
19
GND
C52 Open
-12V/0.25A
GND
C22
+3.3V/2A
TP14
TP15
TP16
SC2453
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SC2453
POWER MANAGEMENT Bill of Materials - Evaluation Board
Item 1 2 3 4 5 6 7 8 9 Quantity 4 1 2 5 1 2 1 1 4 Reference C1,C2,C3,C4 C5 C6,C50 C7,C13,C17,C22,C23 C8 C16 C9 C10 C11 C12 C18 C21 C52 C14 C24 C15 C19 C20 C25 C27 C28 C29 C33 C30 C32 C51 C31 C53 D2 D1 D3 L1 L2 Q4 Q1 Q2 Q3 Q5 Q6 Q7 R1 R13 R24 R51 R2 R3 R4 R12 R5 Part 10uF/16V 10uF 1uF 10uF 0.1uF 0.1uF 0.1uF 0.01uF 680uF/4V Open Open Open Open 8.2nF 680pF 680pF 3.3nF 470uF/6.3V 4.7nF Open Open Open Open 22uF 22uF 22uF 2.2nF 100pF SL04 SL04 B150 STQ127_1042 10uH FZT749 FZT749 SUD50N03-10CP SUD50N03-10CP SUD50N03-10CP SUD50N03-10CP FZT649 10K 10K 10K 10K 10 30K 2 2 11.3K
20
10 11 12 13 14 15
1 2 1 1 1 4
16
3
17 18 19 20 28 29 30 31
1 1 2 1 1 1 2 4
32 33
1 4
34 35 36 37
1 1 2 1
Manufacturer Manufacturer # Foot Print TDK C3216X5R1C106MT SM/C_1206 SM/C_0805 SM/C_0805 TDK C3216X5R1C106MT SM/C_1206 SM/R_0805 SM/C_0805 SM/C_0805 SM/C_0805 SANYO 4TPB680M SM/CT_7343 SM/C_0805 SM/C_0805 SM/C_0805 SM/C_0805 SM/C_0805 SM/C_0805 SM/C_0805 SM/C_0805 SANYO 6TPB470M SM/CT_7343 SM/C_0805 SM/C_1210 SM/C_1210 SM/C_1210 SM/C_1210 SM/C_1210 SM/C_1210 SM/C_1210 SM/C_0805 SM/C_0805 VISAY SL04 SOD123 VISAY SL04 SOD123 VISAY SS15 SM/_SMB DELTA STQ127_1042 COILCRAFT DO5022P-103 DO5022 ZETEX FZT749TA SM/SOT223 ZETEX FZT749TA SM/SOT223 VISHAY SUD50N03-10CP SM/_DPAK VISHAY SUD50N03-10CP SM/_DPAK VISHAY SUD50N03-10CP SM/_DPAK VISHAY SUD50N03-10CP SM/_DPAK ZETEX FZT649 SM/SOT223 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805
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2005 Semtech Corp.
SC2453
POWER MANAGEMENT Bill of Materials - Evaluation Board
38 3 R6 R14 R15 R7 R8 R17 R9 R16 R21 R50 R10 R11 R18 R19 R20 R23 R22 R52 U1 0 0 0 470 1K 1K 20K 20K 20K 20K 2.0K 40.2K 150 2.21K 36K 51K 51K 0 SC2453 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 SM/R_0805 TSSOP-28
39 40 41
1 2 4
42 43 44 45 46 47 48 49
1 1 1 1 1 2 1 1
SEMTECH
SC2453
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SC2453
POWER MANAGEMENT Outline Drawing - TSSOP-28
A e N 2X E/2 E1 PIN 1 INDICATOR ccc C 2X N/2 TIPS 123 E D
DIM
A A1 A2 b c D E1 E e L L1 N 01 aaa bbb ccc
DIMENSIONS INCHES MILLIMETERS MIN NOM MAX MIN NOM MAX
.047 .002 .006 .031 .042 .007 .012 .003 .007 .378 .382 .386 .169 .173 .177 .252 BSC .026 BSC .018 .024 .030 (.039) 28 0 8 .004 .004 .008 1.20 0.05 0.15 0.80 1.05 0.19 0.30 0.09 0.20 9.60 9.70 9.80 4.30 4.40 4.50 6.40 BSC 0.65 BSC 0.45 0.60 0.75 (1.0) 28 0 8 0.10 0.10 0.20
e/2 B D A2 A
aaa C SEATING PLANE
C
bxN bbb
A1 C A-B D GAGE PLANE 0.25
H c
L (L1) DETAIL
01
SIDE VIEW
SEE DETAIL
A
A
NOTES: 1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES). 2. DATUMS -A- AND -B- TO BE DETERMINED AT DATUM PLANE -H3. DIMENSIONS "E1" AND "D" DO NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. 4. REFERENCE JEDEC STD MO-153, VARIATION AE.
Land Pattern - TSSOP-28
X
DIM
(C) G Z C G P X Y Z
DIMENSIONS INCHES MILLIMETERS
(.222) .161 .026 .016 .061 .283 (5.65) 4.10 0.65 0.40 1.55 7.20
Y P
NOTES: 1. THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY. CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR COMPANY'S MANUFACTURING GUIDELINES ARE MET.
Contact Information
Semtech Corporation Power Management Products Division 200 Flynn Road, Camarillo, CA 93012 Phone: (805)498-2111 FAX (805)498-3804
2005 Semtech Corp. 22 www.semtech.com

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