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pin configurations a rail-to-rail high output current operational amplifiers op179/op279 general description the op179 and op279 are rail-to-rail, high output current, single-supply amplifiers. they are designed for low voltage applications that require either current or capacitive load drive capability. the op179/op279 can sink and source currents of 60 ma (typical) and are stable with capacitive loads to 10 nf. applications that benefit from the high output current of the op179/op279 include driving headphones, displays, transform- ers and power transistors. the powerful output is combined with a unique input stage that maintains very low distortion with wide common-mode range, even in single supply designs. the op179/op279 can be used as a buffer to provide much greater drive capability than can usually be provided by cmos outputs. cmos asics and dac often have outputs that can swing to both the positive supply and ground, but cannot drive more than a few milliamps. bandwidth is typically 5 mhz and the slew rate is 3 v/ s, making these amplifiers well suited for single supply applications that require audio bandwidths when used in high gain configurations. operation is guaranteed from voltages as low as 4.5 v, up to 12 v. very good audio performance can be attained when using the op179/op279 in 5 volt systems. thd is below 0.01% with a 600 ? load, and noise is a respectable 21 nv/ hz . supply current is less than 3.5 ma per amplifier. the single op179 is available in the 5-lead sot-23-5 package. it is specified over the industrial (C40 c to +85 c) tempera- ture range. the op279 is available in 8-lead tssop and so-8 surface mount packages. they are specified over the industrial (C40 c to +85 c) temperature range. 8-lead soic and tssop so-8 (s) and ru-8 1 2 3 4 8 7 6 5 op279 in a v +in a out b in b v+ +in b out a features rail-to-rail inputs and outputs high output current: 60 ma single supply: 5 v to 12 v wide bandwidth: 5 mhz high slew rate: 3 v/ s low distortion: 0.01% unity-gain stable no phase reversal short-circuit protected drives capacitive loads: 10 nf applications multimedia telecom daa transformer driver lcd driver low voltage servo control modems fet drivers 5-lead sot-23-5 (rt-5) 1 2 3 5 4 in a +in a v out a op179 v+ 8-lead soic (s suffix) 1 2 3 4 8 7 6 5 op179 in a v +in a v+ out a nc nc nc nc = no connect rev. g information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781/329-4700 www.analog.com fax: 781/326-8703 ? analog devices, inc., 2002
electrical specifications (@ v s = 5.0 v, v cm = 2.5 v, ?0 c t a +85 c unless otherwise noted.) parameter symbol conditions min typ max unit input characteristics offset voltage op179 v os v out = 2.5 v 5mv op279 v os v out = 2.5 v 4mv input bias current i b v out = 2.5 v, t a = 25 c 300 na v out = 2.5 v 700 na input offset current i os v out = 2.5 v, t a = 25 c 50 na v out = 2.5 v 100 na input voltage range v cm 05v common-mode rejection ratio cmrr v cm = 0 v to 5 v 56 66 db large signal voltage gain a vo r l = 1 k ? , 0.3 v v out 4.7 v 20 v/mv offset voltage drift ? v os / ? t4 v/ c output characteristics output voltage high v oh i l = 10 ma source +4.8 v output voltage low v ol i l = 10 ma sink, t a = 25 c75mv i l = 10 ma sink 100 mv short-circuit limit i sc t a = 25 c 40 ma output impedance z out f = 1 mhz, a v = 1 22 ? power supply power supply rejection ratio psrr v s = 4.5 v to 12 v 70 88 db supply current/amplifier i sy v out = 2.5 v 3.5 ma supply voltage range v s +4.5 +12 v dynamic performance slew rate sr r l = 1 k ? , 1 nf 3 v/ s gain bandwidth product gbp 5 mhz phase margin m 60 degrees capacitive load drive no oscillation 10 nf audio performance total harmonic distortion thd 0.01 % voltage noise density e n f = 1 khz 22 nv/ hz electrical specifications (@ v s = 5.0 v, ?0 c t a +85 c unless otherwise noted.) parameter symbol conditions min typ max unit input characteristics offset voltage op179 v os v out = 0 5mv op279 v os v out = 0 4mv input bias current i b t a = 25 c 300 na 700 na input offset current i os t a = 25 c 50 na 100 na input voltage range v cm C5 +5 v common-mode rejection ratio cmrr v cm = C5 v to +5 v 56 66 db large signal voltage gain a vo r l = 1 k ? , C4.7 v v out 4.7 v 20 v/mv offset voltage drift ? v os / ? t3 v/ c output characteristics output voltage high v oh i l = 10 ma source +4.8 v output voltage low v ol i l = 10 ma sink C4.85 v short circuit limit i sc t a = 25 c 50 ma open-loop output impedance z out f = 1 mhz, a v = +1 22 ? power supply supply current/amplifier i sy v s = 6 v, v out = 0 v 3.75 ma dynamic performance slew rate sr r l = 1 k ? , 1 nf 3 v/ s full-power bandwidth bw p 1% distortion khz gain bandwidth product gbp 5 mhz phase margin m 69 degrees noise performance voltage noise e n p-p 0.1 hz to 10 hz 2 v p-p voltage noise density e n f = 1 khz 22 nv/ hz current noise density i n 1 pa/ hz specifications subject to change without notice. op179/op279?pecifications C2C rev. g
op179/op279 C3C rev. g absolute maximum ratings supply voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 v input voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 v differential input voltage 1 . . . . . . . . . . . . . . . . . . . . . . . . . 1 v output short-circuit duration to gnd . . . . . . . . . . indefinite storage temperature range s, rt, ru package . . . . . . . . . . . . . . . . . . C65 c to +150 c operating temperature range op179g/op279g . . . . . . . . . . . . . . . . . . . . C40 c to +85 c junction temperature range s, rt, ru package . . . . . . . . . . . . . . . . . . . C65 c to +150 c lead temperature range (soldering, 60 sec) . . . . . . . . 300 c caution esd (electrostatic discharge) sensitive device. electrostatic charges as high as 4000 v readily accumulate on the human body and test equipment and can discharge without detection. although the op179/op279 features proprietary esd protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. therefore, proper esd precautions are recommended to avoid performance degradation or loss of functionality. ordering guide package temperature range package description package option brand code op179grt C40 c to +85 c 5-lead sot-23 rt-5 a2g op279gs C40 c to +85 c 8-lead soic so-8 op279gru C40 c to +85 c 8-lead tssop ru-8 warning! esd sensitive device package types ja 2 jc unit 5-lead sot-23 (rt) 256 81 c/w 8-lead soic (s) 158 43 c/w 8-lead tssop (ru) 240 43 c/w notes 1 the inputs are clamped with back-to-back diodes. if the differential input voltage exceeds 1 volt, the input current should be limited to 5 ma. 2 ja is specified for the worst-case conditions, i.e., ja is specified for device soldered in circuit board for soic packages.
op179/op279 C4C rev. g 160 0 2.5 40 20 2.5 80 60 100 120 140 1.5 0.5 0.5 1.5 input offset mv units v s 5v t a 25 c 620 op amps tpc 1. input offset distribution common-mode voltage volts offset voltage mv 3.0 0 5 1.5 0.5 1 1.0 0 2.5 2.0 4 3 2 v s 5v t a 25 c tpc 4. offset voltage vs. common-mode voltage 1000 0 100 600 200 25 400 50 800 75 50 25 0 temperature c r l 2k r l 1k open-loop gain v/mv v s 15v 0.3 v out 4.7v tpc 7. open-loop gain vs. temperature v s 5v v cm 2.5v 90 40 100 70 50 25 60 50 80 75 50 25 0 temperature c i sc short-circuit current ma +i sc tpc 2. short-circuit current vs. temperature v s 5v 100 50 100 80 60 25 70 50 90 75 50 25 0 temperature c short-circuit current ma i sc +i sc tpc 5. short-circuit current vs. temperature 5 0 100 3 1 25 2 50 4 75 50 25 0 temperature c slew rate v s v s 5v r l 1k c l +1nf +edge edge tpc 8. slew rate vs. temperature 400 400 5 200 300 1 0 0 100 100 200 300 4 3 2 +85 c +25 c common-mode voltage volts input bias current na v s 5v 40 c tpc 3. input bias current vs. common-mode voltage common-mode voltage volts 7 0 5 3 1 1 2 0 6 4 5 4 3 2 bandwidth mhz v s 5v t a 25 c tpc 6. bandwidth vs. common-mode voltage phase gain 40 40 100 1k 10m 1m 100k 10k 60 80 100 20 0 20 90 90 135 180 225 45 0 45 frequency hz open-loop gain db phase degrees 120 270 v s 2.5v t a 40 c r l 2k tpc 9. open-loop gain and phase vs. frequency typical performance characteristics
op179/op279 C5C rev. g v s 5v v cm 2.5v 6.5 4.0 100 5.5 4.5 25 5.0 50 6.0 75 50 25 0 temperature c supply current ma v s 6v v s 5v tpc 10. supply current vs. temperature frequency hz power supply rejection db 120 60 0 10 100 10m 1m 100k 10k 1k 80 100 20 40 v s 2.5v t a 25 c psrr +psrr tpc 13. power supply rejection vs. frequency 12 6 0 10k 10m 1m 100k 1k 4 2 8 10 frequency hz maximum output swing volts t a 25 c v s 5v a vcl +1 r l 1k tpc 16. maximum output swing vs. frequency 5 0 100 3 1 25 2 50 4 75 50 25 0 temperature c slew rate v/ s v s 5v r l 1k c l +1nf +edge edge tpc 11. slew rate vs. temperature 6 3 0 10k 10m 1m 100k 1k 2 1 4 5 frequency hz maximum output swing volts t a 25 c v s 2.5v a vcl +1 r l 1k tpc 14. maximum output swing vs. frequency 50 10 30 1k 10k 100m 10m 1m 100k 20 30 40 20 10 0 frequency hz closed-loop gain db a vcl +100 a vcl +10 a vcl +1 v s 2.5v t a 25 c r l 1k tpc 17. closed-loop gain vs. frequency phase gain 120 40 40 100 1k 10m 1m 100k 10k 60 80 100 20 0 20 270 90 90 135 180 225 45 0 45 frequency hz open-loop gain db phase degrees v s 2.5v t a 40 c r l 2k c l 500pf tpc 12. open-loop gain and phase vs. frequency 180 160 140 120 100 80 60 40 20 0 10 100 10m 1m 100k 10k 1k t a 25 c v s 2.5v or 5v frequency hz impedance a vcl 1 a vcl 10 or 100 &'()0 ( +. * 5
6 80 0 10k 20 10 0 40 30 50 60 70 8k 6k 4k 2k load capacitance pf overshoot % t a 25 c a vcl +1 r l 1k v s 2.5v v in 100mv p-p positive edge and negative edge tpc 18. small signal overshoot vs. load capacitance
op179/op279 C6C rev. g theory of operation the op179/op279 is the latest entry in analog devices expand- ing family of single-supply devices, designed for the multimedia and telecom marketplaces. it is a high output current drive, rail-to-rail input /output operational amplifier, powered from a single 5 v supply. it is also intended for other low supply voltage applications where low distortion and high output current drive are needed. to combine the attributes of high output current and low distortion in rail-to-rail input/output operation, novel circuit design techniques are used. for example, tpc 1 illustrates a simplified equivalent circuit for the op179/op279s input stage. it is comprised of two pnp differential pairs, q5-q6 and q7-q8, operating in parallel, with diode protection networks. diode networks d5-d6 and d7-d8 serve to clamp the applied differential input voltage to the op179/op279, thereby protecting the input transistors against avalanche damage. the fundamental differences between these two pnp gain stages are that the q7-q8 pair are normally off and that their inputs are buffered from the operational amplifier inputs by q1-d1-d2 and q9-d3-d4. operation is best under- stood as a function of the applied common-mode voltage: when the inputs of the op179/op279 are biased midway between the supplies, the differential signal path gain is controlled by the resistively loaded (via r7, r8) q5-q6. as the input common-mode level is reduced toward the negative supply (v neg or gnd), the input transistor current sources, i1 and i3, are forced into satura- tion, thereby forcing the q1-d1-d2 and q9-d3-d4 networks into cutoff; however, q5-q6 remain active, providing input stage gain. on the other hand, when the common-mode input voltage is increased toward the positive supply, q5-q6 are driven into cutoff, q3 is driven into saturation, and q4 becomes active, providing bias to the q7-q8 differential pair. the point at which the q7-q8 differential pair becomes active is approximately equal to (v pos C 1 v). i2 r5 4k d7 i1 r6 4k d8 d5 d6 r3 2.5k r4 2.5k q4 q3 q2 q5 q6 q9 q1 r1 6k r2 3k v pos v neg r7 2.2k r8 2.2k i3 d1 d2 d3 d4 v o + in in+ q8 q7 figure 1. op179/op279 equivalent input circuit the key issue here is the behavior of the input bias currents in this stage. the input bias currents of the op179/op279 over the range of common-mode voltages from (v neg + 1 v) to (v pos C 1 v) are the arithmetic sum of the base currents in q1-q5 and q9-q6. outside of this range, the input bias currents are dominated by the base current sum of q5-q6 for input signals close to v neg , and of q1-q5 (q9-q6) for input signals close to v pos . as a result of this design approach, the input bias currents in the op179/op279 not only exhibit different amplitudes, but also exhibit different polarities. this input bias current behavior is best illustrated in tpc 3. it is, therefore, of paramount importance that the effective source impedances connected to the op179/op279s inputs are balanced for optimum dc and ac performance. 100 60 0 10 10k 1k 100 1 40 20 80 frequency hz voltage noise density nv/ hz v s 5v t a 25 c tpc 19. voltage noise density vs. frequency 120 60 0 1k 1m 100k 10k 100 40 20 80 100 frequency hz common-mode rejection db t a 25 c v s 2.5v tpc 21. common-mode rejection vs. frequency common-mode voltage volts 60 0 5 30 10 1 20 0 50 40 4 3 2 voltage noise density nv/ hz v s 5v t a 25 c frequency 1khz tpc 20. voltage noise density vs. common-mode voltage
op179/op279 C7C rev. g in order to achieve rail-to-rail output behavior, the op179/op279 design employs a complementary common-emitter (or g m r l ) output stage (q15-q16), as illustrated in figure 2. these amplifiers provide output current until they are forced into saturation, which occurs at approximately 50 mv from either supply rail. thus, their saturation voltage is the limit on the maximum output voltage swing in the op179/op279. the output stage also exhibits voltage gain, by virtue of the use of common-emitter amplifiers; and, as a result, the voltage gain of the output stage (thus, the open-loop gain of the device) exhib- its a strong dependence to the total load resistance at the output of the op179/op279 as illustrated in tpc 7. q7 q3 q15 q9 105 v pos v neg q13 v out q4 q16 i3 i4 q11 q12 q5 q10 i2 q1 q2 i1 q8 q6 105 q14 150 figure 2. op179/op279 equivalent output circuit input overvoltage protection as with any semiconductor device, whenever the condition exists for the input to exceed either supply voltage, the devices input overvoltage characteristic must be considered. when an overvoltage occurs, the amplifier could be damaged, depending on the magnitude of the applied voltage and the magnitude of the fault current. figure 3 illustrates the input overvoltage char- acteristic of the op179/op279. this graph was generated with the power supplies at ground and a curve tracer connected to the input. as can be seen, when the input voltage exceeds either supply by more than 0.6 v, internal pn-junctions energize, which allows current to flow from the input to the supplies. as illustrated in the simplified equivalent input circuit (figure 1), the op179/op279 does not have any internal current limiting resistors, so fault currents can quickly rise to damaging levels. this input current is not inherently damaging to the device as long as it is limited to 5 ma or less. for the op179/op279, once the input voltage exceeds the supply by more than 0.6 v, the input current quickly exceeds 5 ma. if this condition continues to exist, an external series resistor should be added. the size of the resistor is calculated by dividing the maximum overvoltage by 5 ma. for example, if the input voltage could reach 100 v, the external resistor should be (100 v/5 ma) = 20 k ? . this resis- tance should be placed in series with either or both inputs if they are exposed to an overvoltage. again, in order to ensure optimum dc and ac performance, it is important to balance source imped- ance levels. for more information on general overvoltage charac- teristics of amplifiers refer to the 1993 seminar applications guide, available from the analog devices literature center. 5 3 5 2.0 4 1 2 1 2 3 4 2.0 1.0 0 1.0 0 input current ma input voltage v figure 3. op179/op279 input overvoltage characteristic output phase reversal some operational amplifiers designed for single-supply operation exhibit an output voltage phase reversal when their inputs are driven beyond their useful common-mode range. typically for single-supply bipolar op amps, the negative supply determines the lower limit of their common-mode range. with these devices, external clamping diodes, with the anode connected to ground and the cathode to the inputs, input signal excursions are pre- vented from exceeding the devices negative supply (i.e., gnd), preventing a condition that could cause the output voltage to change phase. jfet input amplifiers may also exhibit phase reversal and, if so, a series input resistor is usually required to prevent it. the op179/op279 is free from reasonable input voltage range restrictions provided that input voltages no greater than the supply voltages are applied. although the devices output will not change phase, large currents can flow through the input protection diodes, shown in figure 1. therefore, the technique recommended in the input overvoltage protection section should be applied in those applications where the li kelihood of input voltages exceeding the supply voltages is possible. capacitive load drive the op179/op279 has excellent capacitive load driving capa- bilities. it can drive up to 10 nf directly as the performance graph titled small signal overshoot vs. load capacitance (tpc 18) shows. however, even though the device is stable, a capacitive load does not come without a penalty in bandwidth. as shown in figure 4, the bandwidth is reduced to under 1 mhz for loads greater than 3 nf. a snubber network on the output will not increase the bandwidth, but it does significantly reduce the amount of overshoot for a given capacitive load. a snubber consists of a series r-c network (r s , c s ), as shown in figure 5, connected from the output of the device to ground. this net- work operates in parallel with the load capacit or, c l , to provide phase lag compensation. the actual value of the resistor and capacitor is best determined empirically.
op179/op279 C8C rev. g 7 2 0 0.01 0.100 10 1 5 1 3 4 6 capacitive load nf bandwidth mhz v s 5v r l 1k t a 25 c figure 4. op179/op279 bandwidth vs. capacitive load 1/2 op279 r s 20v c s 1 f c l 10nf 5v v in 100mv p-p v out figure 5. snubber network compensates for capacitive load the first step is to determine the value of the resistor, r s . a good starting value is 100 ? (typically, the optimum value will be less than 100 ? ). this value is reduced until the small-signal transient response is optimized. next, c s is determined10 f is a good s tarting point. this value is reduced to the smallest value for acceptable performance (typically, 1 f). for the case of a 10 nf load capacitor on the op179/op279, the optimal snubber network is a 20 ? in series with 1 f. the benefit is immediately apparent as seen in the scope photo in figure 6. the top trace was taken with a 10 nf load and the bottom trace with the 20 ? , 1 f snubber network in place. the amount of overshot and ringing is dramatically reduced. table i illustrates a few sample snubber networks for large load capacitors. 90 100 10nf load only snubber in circuit 10 0% 50mv 2 s figure 6. overshoot and ringing are reduced by adding a snubber network in parallel with the 10 nf load table i. snubber networks for large capacitive loads load capacitance (c l ) snubber network (r s , c s ) 10 nf 20 ? , 1 f 100 nf 5 ? , 10 f 1 f0 ? , 10 f overload recovery time overload, or overdrive, recovery time of an operational amplifier is the time required for the output voltage to recover to its linear region from a saturated condition. this recovery time is impor- tant in applications where the amplifier must recover after a large transient event. the circuit in figure 7 was used to evaluate the op179/op279s overload recovery time. the op179/op279 takes approximately 1 s to recover from positive saturation and approximately 1.2 s to recover from negative saturation. 1/2 op279 r l 499 +5v v out 5v r3 10k r2 1k r1 909 2v p-p @ 100hz figure 7. overload recovery time test circuit output transient current recovery in many applications, operational amplifiers are used to provide moderate levels of output current to drive the inputs of adcs, small motors, transmission lines and current sources. it is in these applications that operational amplifiers must recover quickly to step changes in the load current while maintaining steady-state load current levels. because of its high output current capability and low closed-loop output impedance, the op179/op279 is an excellent choice for these types of applications. for example, when sourcing or sinking a 25 ma steady-state load current, the op179/op279 exhibits a recovery time of less than 500 ns to 0.1% for a 10 ma (i.e., 25 ma to 35 ma and 35 ma to 25 ma) step change in load current. a precision negative voltage reference in many data acquisition applications, the need for a precision negative reference is required. in general, any positive voltage reference can be converted into a negative voltage reference through the use of an operational amplifier and a pair of matched resistors in an inverting configuration. the disadvantage to that approach is that the largest single source of error in the circuit is the relative matching of the resistors used. the circuit illustrated in figure 8 avoids the need for tightly matched resistors with the use of an active integrator circuit. in this circuit, the output of the voltage reference provides the input drive for the integrator. the integrator, to maintain circuit equilibrium, adjusts its output to establish the proper relation- ship between the references v out and gnd. thus, various negative output voltages can be chosen simply by substituting for the appropriate reference ic (see table). to speed up the
op179/op279 C9C rev. g on-off settling time of the circuit, r2 can be reduced to 50 k ? or less. although the integrators time constant chosen here is 1 ms, room exists to trade off circuit bandwidth and noise by increasing r3 and decreasing c2. the shutdown feature is maintained in the circuit with the simple addition of a pnp transistor and a 10 k ? resistor. one caveat with this approach should be mentioned: although rail-to-rail output amplifiers work best in the application, these operational ampli- fiers require a finite amount (mv) of headroom when required to provide any load current. the choice for the circuits negative supply should take this issue into account. r4 10 1/2 op279 +5v 10v r3 1k c2 1 f c1 1 f r2 100k u1 ref195 gnd r5 10k r1 10k 2n3904 4 6 2 3 shutdown ttl/cmos +5v v ref u1 ref192 ref193 ref196 ref194 v out (v) 2.5 3.0 3.3 4.5 figure 8. a negative precision voltage reference that uses no precision resistors exhibits high output current drive a high output current, buffered reference/regulator many applications require stable voltage outputs relatively close in potential to an unregulated input source. this low dropout type of reference/regulator is readily implemented with a rail-to- rail output op amp, and is particularly useful when using a higher current device such as the op179/op279. a typical example is the 3.3 v or 4.5 v reference voltage developed from a 5 v system source. generating these voltages requires a three- terminal reference, such as the ref196 (3.3 v) or the ref194 (4.5 v), both of which feature low power, with sourcing outputs of 30 ma or less. figure 9 shows how such a reference can be outfitted with an op179/op279 buffer for higher currents and/ or voltage levels, plus sink and source load capability. c2 0.1 f r2 10k 1% u2 1/2 op279 v out1 = 3.3v @ 30ma r5 1 c5 10 f/25v tantalum r1 10k 1% c1 0.1 f v s 5v v out2 = 3.3v c4 1 f 6 2 3 4 v out common c3 0.1 f v c on/off control input cmos hi (or open) = on lo = off v s common r3 (see text) r4 3.3k u1 ref196 figure 9. a high output current reference/regulator the low dropout performance of this circuit is provided by stage u2, one-half of an op179/op279 connected as a follower/buffer for the basic reference voltage produced by u1. the low voltage saturation characteristic of the op179/op279 allows up to 30 ma of load current in the illustrated use, as a 5 v to 3.3 v converter with high dc accuracy. in fact, the dc output voltage change for a 30 ma load current delta measures less than 1 mv. this corresponds to an equivalent output impedance of < 0.03 ? . in this application, the stable 3.3 v from u1 is applied to u2 through a noise filter, r1-c1. u2 replicates the u1 voltage within a few mv, but at a higher current output at v out1 , with the ability to both sink and source output current(s)unlike most ic references. r2 and c2 in the feedback path of u2 provide bias compensation for lowest dc error and additional noise filtering. transient performance of the reference/regulator for a 10 ma step change in load current is also quite good and is determined largely by the r5-c5 output network. with values as shown, the transient is about 10 mv peak and settles to within 2 mv in 8 s, for either polarity. although room exists for optimizing the transient response, any changes to the r5-c5 network should be verified by experiment to preclude the possibility of excessive ringing with some capacitor types. to scale v out2 to another (higher) output level, the optional resistor r3 (shown dotted) is added, causing the new v out1 to become: vv r r out1 out2 =+ ? ? ? ? ? ? 1 2 3 as an example, for a v out1 = 4.5 v, and v out2 = 2.5 v from a ref192, the gain required of u2 is 1.8 times, so r2 and r3 would be chosen for a ratio of 0.8:1, or 18 k ? :22.5 k ? . note that for the lowest v out1 dc error, the parallel combination of r2 and r3 should be maintained equal to r1 (as here), and the r2-r3 resistors should be stable, close tolerance metal film types. the circuit can be used as shown as either a 5 v to 3.3 v reference/ regulator, or it can be used with on/off control. by driving pin 3 of u1 with a logic control signal as noted, the output is switched on/off. note that when on/off control is used, resistor r4 should be used with u1 to speed on-off switching. direct access arrangement for telephone line interface figure 10 illustrates a 5 v only transmit/receive telephone line interface for 110 ? transmission systems. it allows full duplex transmission of signals on a transformer coupled 110 ? line in a differential manner. amplifier a1 provides gain that can be adjusted to meet the modem output drive requirements. both a1 and a2 are configured to apply the largest possible signal on a single supply to the transformer. because of the op179/op279s high output current drive and low dropout voltage, the largest signal available on a single 5 v supply is approximately 4.5 v p-p into a 110 ? transmission system. amplifier a3 is configured as a difference amplifier to extract the receive signal from the transmission line for amplification by a4. a4s gain can be adjusted in the same manner as a1s to meet the modems input signal requirements. standard resistor values permit the use of sip (single in-line package) format resistor arrays. couple this with the op179/op279s 8-lead soic footprint and this circuit offers a compact, cost-sensitive solution.
op179/op279 C10C rev. g 6.2v 6.2v transmit txa receive rxa c1 0.1 f r1 10k r2 9.09k 2k p1 tx gain adjust a1 a2 a3 a4 a1, a2 = 1/2 op279 a3, a4 = 1/2 op279 r3 55 r4 55 1:1 t1 to telephone line 1 2 3 7 6 5 2 3 1 6 5 7 10 f r7 10k r8 10k r5 10k r6 10k r9 10k r14 9.09k r10 10k r11 10k r12 10k r13 10k c2 0.1 f p2 rx gain adjust 2k z o 110 5v dc figure 10. a single-supply direct access arrangement for modems a single-supply, remote strain gage signal conditioner the circuit in figure 11 illustrates a way by which the op179/ op279 can be used in a 12 v single supply, 350 ? strain gage signal conditioning circuit. in this circuit, the op179/op279 serves two functions: (1) by servoing the output of the ref43s 2.5 v output across r1, it provides a 20 ma drive to the 350 ? strain gage. in this way, small changes in the strain gage pro- duce large differential output voltages across the amp04s inputs. (2) to maximize the circuits dynamic range, the other half of the op179/op279 is configured as a supply-splitter connected to the amp04s ref terminal. thus, tension or compression in the application can be measured by the circuit. ref43 amp04 0.1 f 2 6 4 2.5v 3 1 8 4 2 a1 7 1 8 6 3 2 4 c x c2 0.1 f r4 1k 12v 5 v o 80mv/ v o common r1 124 0.1%, low tcr 100-ft twisted pair belden type 9502 s+ s 350 strain gage f f+ a2 12v r2 10k r3 10k c1 10 f 7 6 5 +6v a1, a2 = 1/2 op279 12v 20ma drive figure 11. a single-supply, remote strain gage signal conditioner the amp04 is configured for a gain of 100, producing a circuit sensitivity of 80 mv/ ? . capacitor c2 is used across the amp04s pins 8 and 6 to provide a 16-hz noise filter. if additional noise filtering is required, an optional capacitor, c x , can be used across the amp04s input to provide differential-mode noise rejection. a single-supply, balanced line driver the circuit in figure 12 is a unique line driver circuit topology used in professional audio applications and has been modified for automotive audio applications. on a single 12 v supply, the line driver exhibits less than 0.02% distortion into a 600 ? load across the entire audio band (not shown). for loads greater than 600 ? , distortion performance improves to where the circuit exhibits less than 0.002%. the design is a transformerless, balanced transmission system where output common-mode rejection of noise is of paramount importance. like the transformer-based system, either output can be shorted to ground for unbalanced line driver applications without changing the circuit gain of 1. other circuit gains can be set according to the equation in the diagram. this allows the design to be easily configured for noninverting, inverting, or differential operation. r l 600 c1 22 f a2 7 6 5 3 1 2 a1 12v r1 10k r2 10k r11 10k r7 10k 6 7 5 a1 12v 12v r8 100k r9 100k c2 1 f r12 10k r14 50 a2 1 2 3 r3 10k r6 10k r13 10k c3 47 f v o1 v o2 c4 47 f a1, a2 = 1/2 op279 gain = r3 r2 set: r7, r10, r11 = r2 set: r6, r12, r13 = r3 v in r5 50 figure 12. a single-supply, balanced line driver for automotive applications
op179/op279 C11C rev. g unity-gain, sallen-key (vcvs) filters high pass configurations figure 14a is the hp form of a unity-gain 2-pole sk filter using an op179/op279 section. for this filter and its lp coun- terpart, the gain in the passband is inherently unity, and the signal phase is noninverting due to the follower hookup. for simplicity and practicality, capacitors c1-c2 are set equal, and resistors r2-r1 are adjusted to a ratio n, which provides the filter damping as per the design expressions. an hp design starts with the selection of standard capacitor values for c1 and c2, and a calculation of n. r1 and r2 are then calculated as per the figure expressions. in these examples, (or 1/q) is set equal to 2 , providing a butterworth (maximally flat) response characteristic. the filter corner frequency is normalized to 1 khz, with resistor values shown in both rounded and (exact) form. various other two-pole response shapes are also possible with appropriate selection of . for a given response type ( ), frequency can be easily scaled, using proportional r or c values. +v s v s u1a op279 1 3 2 4 8 in r2 22k (22.508k ) r1 11k (11.254k ) c2 0.01 f r = r2 0.1 f z f (high pass) c1 0.01 f given: alpha, f set c1 = c2 = c alpha = 2/(n^0.5) = 1/q n = 4/(alpha)^2 = r2/r1 r1 = 1/(2 * pi * f * c * (n^0.5)) r2 = n * r1 1khz bw shown out 7 5 6 r = r1+r2 z f (low pass) given: alpha, f set r1 = r2 = r alpha = 2/(m^0.5) = 1/q n = 4/(alpha)^2 = c2/c1 pick c1 c1 = m*c1 r = 1/(2*p1*f*c1* (m^0.5)) 1khz bw shown in r2 11k (11.254k ) c2 0.01 f 0.1 f c1 0.02 f out u1b op279 r1 11k (11.254k ) a. high pass b. low pass figure 14. two-pole unity-gain sallen key hp/lp filters low pass configurations in the lp sk arrangement of figure 14b, r and c elements are interchanged, and the resistors are made equal. here the c2/c1 ratio m is used to set the filter , as noted. this design is begun with the choice of a standard capacitor value for c1 and a calcu- lation of m, which forces a value of m c1 for c2. then, the value r for r1 and r2 is calculated as per the expression. for highest performance, the passive components used for tun- ing active filters deserve attention. resistors should be 1%, low tc, metal film types of the rn55 or rn60 style, or similar. a single-supply headphone amplifier because of its high speed and large output drive, the op179/p279 makes for an excellent headphone driver, as illustrated in figure 13. its low supply operation and rail-to-rail inputs and outputs give a maximum signal swing on a single 5 v supply. to ensure maximum signal swing available to drive the headphone, the amplifier inputs are biased to v+/2, which is in this case 2.5 v. the 100 k ? resistor to the positive supply is equally split into two 50 k ? with their common point bypassed by 10 f to pre- vent power supply noise from contaminating the audio signal. 16 50k 220 f left headphone 10 f 50k 50k 100k 10 f left input +v + 5v 1/2 op279 16 50k 220 f right headphone 10 f 50k 50k 100k 10 f right input +v +v + 5v 1/2 op279 figure 13. a single-supply, stereo headphone driver the audio signal is then ac-coupled to each input through a 10 f capacitor. a large value is needed to ensure that the 20 hz audio information is not blocked. if the input already has the proper dc bias, the ac coupling and biasing resistors are not required. a 220 f capacitor is used at the output to couple the amplifier to the headphone. this value is much larger than that used for the input because of the low impedance of the head- phones, which can range from 32 ? to 600 ? . an additional 16 ? resistor is used in series with the output capacitor to pro- tect the op amps output stage by limiting capacitor discharge current. when driving a 48 ? load, the circuit exhibits l ess than 0.02% thd+n at low output drive levels (not shown). the op179/op 279s high current output stage can drive this heavy load to 4 v p-p and maintain less than 1% thd+n. active filters several active filter topologies are useful with the op179/op279. among these are two popular architectures, the familiar sallen- key (sk) voltage controlled voltage source (vcvs) and the multiple feedback (mfb) topologies. these filter types can be arranged for high pass (hp), low pass (lp), and band-pass (bp) filters. the sk filter type uses the op amp as a fixed gain voltage follower at unity or a higher gain, while the mfb structure uses it as an inverting stage. discussed here are simplified, 2-pole forms of these filters, highly useful as system building blocks.
op179/op279 C12C rev. g loading can be tempered somewhat by using a small series input resistance of about 100 ? , but can still be an issue. 7 6 5 0.1 f given: alpha, f and h (passband gain) alpha = 1/q pick a std c1 value, then: c3 = c1, c2 = c1/h r1 = alpha/((2 * pi * f * c1) * (2+(1/h))) r2 = (h * (2+(1/h)))/(alpha * (2 * pi * f * c1)) 1khz bw example shown (note: see text on c1 loading considerations) in r1 7.5k out u1b op279 r2 33.6k c3 0.01 f c2 0.01 f c1 0.01 f z b r = r2 figure 15. two-pole, high pass multiple feedback filters in this example, the filter gain is set to unity, the corner fre- quency is 1 khz, and the response is a butterworth type. for applications where dc output offset is critical, bias current com- pensation can be used for the amplifier. this is provided by network z b , where r is equal to r2, and the capacitor provides a noise bypass. low pass configurations figure 16 is a lp mfb 2-pole filter using an op179/op279 section. for this filter, the gain in the pass band is user con- figurable over a wide range, and the pass band signal phase is inverting. given the design parameters for , f, and h, a simplified design process is begun by picking a standard value for c2. then c1 and resistors r1-r3 are selected as per the relationships noted. optional dc bias current compensation is provided by z b , where r is equal to the value of r3 plus the parallel equivalent value of r1 and r2. 7 5 6 (r1 r2)+r3 given: alpha, f and h (passband gain) alpha = 1/q pick a std c2 value, then: c1 = c2 (4 (h +1))/alpha^2 r1 = alpha/(4 h pi f c2) r2 = h r1 r3 = alpha/(4 (h + 1) pi f c2) 1khz bw example shown (note: see text on c1 loading considerations) in out u1b op279 r1 11.3k r2 11.3k r3 5.62k c2 0.01 f 0.1 f z b c1 0.04 f figure 16. two-pole, low-pass multiple feedback filters gain of this filter, h, is set here by resistors r2 and r1 (as in a standard op amp inverter), and can be just as precise as these resistors allow at low frequencies. because of this flexible and accurate gain characteristic, plus a low range of component value spread, this filter is perhaps the most practical of all the mfb types. capacitor ratios are best satisfied by paralleling two or more common types, as in the example, which is a 1 khz unity-gain butterworth filter. capacitors should be 1% or 2% film types preferably, such as polypropylene or polystyrene, or npo (cog) ceramic for smaller values. somewhat lesser performance is available with the use of polyester capacitors. parasitic effects in sallen-key implementations in designing these circuits, moderately low (10 k ? or less) val- ues for r1-r2 can be used to minimize the effects of johnson noise when critical, with, of course, practical tradeoffs of capaci- tor size and expense. dc errors will result for larger values of resistance, unless bias current compensation is used. to add bias compensation in the hp filter of figure 14a, a feedback compensation resistor with a value equal to r2 is used, shown optionally as z f . this will minimize bias induced offset, reduc- ing it to the product of the op179/op279s i os and r2. similar compensation is applied to the lp filter, using a z f resistance of r1 + r2. using dc compensation and relatively low filter values, filter output dc errors using the op179/op279 will be domi- nated by v os , which is limited to 4 mv or less. a caveat here is that the additional resistors increase noise substantiallyfor example, an unbypassed 10 k ? resistor generates 12 nv/ hz of noise. however, the resistance can be ac-bypassed to elimi- nate noise with a simple shunt capacitor, such as 0.1 f. sallen-key implementations in single-supply applications the hookups shown illustrate a classical dual supply op amp application, which for the op179/op279 would use supplies up to 5 v. however, these filters can also use the op amp in a single-supply mode, with little if any alteration to the filter itself. to operate single supply, the op179/op279 is powered from 5 v at pin 8 with pin 4 grounded. the input dc bias for the op amp must be supplied from a dc source equal to one-half supply, or 2.5 v in this case. for the hp section, dc bias is applied to the common end of r2. r2 is simply returned to an ac ground that is a well-bypassed 2:1 divider across the 5 v source. this can be as simple as a pair of 100 k ? resistors with a 10 f bypass cap. the output from the stage is then ac coupled, using an appropriate coupling cap from u1a to the next stage. for the lp section dc bias is applied to the input end of r1, in common with the input signal. this dc can be taken from an unbypassed dual 100 k ? divider across the supply, with the input signal ac coupled to the divider and r1. multiple feedback filters mfb filters, like their sk relatives, can be used as building blocks as well. they feature lp and hp operation as well, but can also be used in a band-pass bp mode. they have the property of inverting operation in the pass band, since they are based on an inverting amplifier structure. another useful asset is their ability to be easily configured for gain. high pass configurations figure 15 shows an hp mfb 2-pole filter using an op179/ op279 section. for this filter, the gain in the pass band is user configurable, and the signal phase is inverting. the circuit uses one more tuning component than the sk types. for simplicity, capacitors c1 and c3 are set to equal standard values, and resis- tors r1-r2 are selected as per the relationships noted. gain of this filter, h, is set by capacitors c1 and c2, and this factor limits both gain selectability and precision. also, input capaci- tance c1 makes the load seen by the driving stage highly reactive, and limits overall practicality of this filter. the dire effect of c1
op179/op279 C13C rev. g v in 3 2 1 u1a op279 +v s 4 v s r1 31.6k c1 0.01 f c2 0.01 f r2 31.6k r5 31.6k r6 31.6k r4 49.9 hi lo 500hz and up dc 500hz 6 5 7 c3 0.01 f u1b op279 c4 0.02 f r7 15.8k r3 49.9 0.1 f 0.1 f 100 f/25v 100 f/25v +v s v s to u1 +5v 5v com figure 18. two-way active crossover networks in the filter sections, component values have been selected for good balance between reasonable physical/electrical size, and lowest noise and distortion. dc offset errors can be minimized by using dc compensation in the feedback and bias paths, ac bypassed with capacitors for low noise. also, since the network input is reactive, it should driven from a directly coupled low impedance source at v in . figure 19 shows this filter architecture adapted for single-supply operation from a 5 v dc source, along the lines discussed previously. v in 3 2 1 u1a op279 +v s 4 r1 31.6k c1 0.01 f c2 0.01 f r2 31.6k r5 31.6k r6 31.6k r4 49.9 hi lo 500hz and up dc 500hz 6 5 7 c3 0.01 f u1b op279 c4 0.02 f r7 15.8k r3 49.9 10 f 10 f 100k +v s 10 f 100k 100k c in 10 f r in 100k 0.1 f 100 f/25v +v s to u1 +5v com + 100k + figure 19. a single-supply, two-way active crossover band-pass configurations the mfb band-pass filter using an op179/op279 section is shown in figure 17. this filter provides reasonably stable medium q designs for frequencies of up to a few khz. for best pre- dictability and stability, operation should be restricted to applications where the op179/op279 has an open-loop gain in excess of 2q 2 at the filter center frequency. 7 6 5 r = r3 0.1 f given: q, f, and a o (passband gain) alpha = 1/q, h = a o /q pick a std c1 value, then: c2 = c1 r1 = 1/(h*(2*pi*f*c1)) r2 = 1/(((2*q) h)*(2*pi*f*c1)) r3 = q/(pi*f*c1) example: 60hz, q = 10, a o = 10 (or 1) a o = 1 for '( )' values in r2 1.4k (1.33k ) out u1b op279 r3 530k c2 0.1 f c1 0.1 f z b r1 26.4k (264k ) figure 17. two-pole, band-pass multiple feedback filters given the band-pass design parameters for q, f, and pass band gain a o , the design process is begun by picking a standard value for c1. then c2 and resistors r1-r3 are selected as per the relationships noted. this filter is subject to a wide range of component values by nature. practical designs should attempt to restrict resistances to a 1 k ? to 1 m ? range, with capacitor values of 1 f or less. when needed, dc bias current compensa- tion is provided by z b , where r is equal to r3. two-way loudspeaker crossover networks active filters are useful in loudspeaker crossover networks for reasons of small size, relative freedom from parasitic effects, and the ease of controlling low/high channel drive, plus the con- trolled driver damping provided by a dedicated amplifier. both sallen-key (sk) vcvs and multiple-feedback (mfb) filter architectures are useful in implementing active crossover networks (see reference 4, page 14 ), and the circuit shown in figure 18 is a two-way active crossover that combines the advan- tages of both filter topologies. this active crossover exhibits less than 0.01% thd+n at output levels of 1 v rms using general purpose unity gain hp/lp stages. in this two-way example, the lo signal is a dc-500 hz lp woofer output, and the hi signal is the hp (> 500 hz) tweeter output. u1b forms an mfb lp section at 500 hz, while u1a provides an sk hp section, cov- ering frequencies 500 hz. this crossover network is a linkwitz-riley type (see reference 5, page 14 ), with a damping factor or of 2 (also referred to as butterworth squared). a hallmark of the linkwitz-riley type of filter is the fact that the summed magnitude response is flat across the pass band. a necessary condition for this to happen is the relative signal polarity of the hi output must be inverted with respect to the low outputs. if only sk filter sections were used, this requires that the connections to one speaker be reversed on installation. alternately, with one inverting stage used in the lo channel, this accomplishes the same effect. in the circuit as shown, stage u1b is the mfb lp filter, which provides the necessary polarity inversion. like the sk sections, it is config- ured for unity gain and an of 2. the cutoff frequency is 500 hz, which complements the sk hp section of u4.
op179/op279 C14C rev. g references on active filters and active crossover networks 1. sallen, r.p.; key, e.l., a practical method of designing rc active filters, ire transactions on circuit theory , vol. ct-2, march 1955. 2. huelsman, l.p.; allen, p.e., introduction to the theory and design of active filters , mcgraw-hill, 1980. 3. zumbahlen, h., chapter 6: passive and active analog filtering, within 1992 analog devices amplifier applications guide . 4. zumbahlen, h., speaker crossovers, within chapter 8 of 1993 analog devices system applications guide. 5. linkwitz, s., active crossover networks for noncoincident drivers, j aes , vol. 24, #1, jan/feb 1976. the crossover example frequency of 500 hz can be shifted lower or higher by frequency scaling of either resistors or capacitors. in configuring the circuit for other frequencies, complementary lp/ hp action must be maintained between sections, and component values within the sections must be in the same ratio. table ii provides a design aid to adaptation, with suggested standard component values for other frequencies. table ii. rc component selection for various crossover frequencies r1/c1 (u1a) * crossover frequency (hz) r5/c3 (u1b) ** 100 160 k ? /0.01 f 200 80.6 k ? /0.01 f 319 49.9 k ? /0.01 f 500 31.6 k ? /0.01 f 1 k 16 k ? /0.01 f 2 k 8.06 k ? /0.01 f 5 k 3.16 k ? /0.01 f 10 k 1.6 k ? /0.01 f table notes (applicable for = 2). * * for sk stage u1a: r1 = r2, and c1 = c2, etc. ** for mfb stage u1b: r6 = r5, r7 = r5/2, and c4 = 2c3.
op179/op279 C15C rev. g r10 16 98 10 c3 15 16 15.915e-12 * * zero at 1.5 mhz * e1 14 98 (9,39) 1e6 r5 14 18 1e6 r6 18 98 1 c4 14 18 106.103e-15 * * bias current-vs-common-mode voltage * ep 97 0 (99,0) 1 en 51 0 (50,0) 1 v3 20 21 1.6 v4 22 23 2.8 r12 97 20 530 r13 23 51 1e3 d13 15 21 dx d14 22 15 dx fib 98 24 poly(2) v3 v4 0 C1 1 rib 24 98 10e3 e3 97 25 poly(1) (99,39) C1.63 1 e4 26 51 poly(1) (39,50) C2.73 1 d15 24 25 dx d16 26 24 dx * * pole at 6 mhz * g6 98 40 (18,39) 1e 6 r20 40 98 1e6 c10 40 98 26.526e-15 * * output stage * rs1 99 39 6.0345e3 rs2 39 50 6.0345e3 ro1 99 45 40 ro2 45 50 40 g7 45 99 (99,40) 25e-3 g8 50 45 (40,50) 25e-3 g9 98 60 (45,40) 25e-3 d9 60 61 dx d10 62 60 dx v7 61 98 dc 0 v8 98 62 dc 0 fsy 99 50 poly(2) v7 v8 1.711e-3 1 1 d11 41 45 dz d12 45 42 dz v5 40 41 1.54 v6 42 40 1.54 .model dx d() .model dz d(is=1e-6) .model qn npn(bf=300) .ends op179/op279 spice macro model * op179/op279 spice macro model rev. a, 5/94 * arg / adi * * copyright 1994 by analog devices * * refer to readme.doc file for license statement. use of * this model indicates your acceptance of the terms and pro- * visions in the license statement. * * node assignments * noninverting input * | inverting input * | | positive supply * | | | negative supply * |||| output * ||||| .subckt op179/op279 3 2 99 50 45 * * input stage and pole at 6 mhz * i1 1 50 60.2e-6 q1 5 2 7 qn q2 6 4 8 qn d1 4 2 dx d2 2 4 dx r1 1 7 1.628e3 r2 1 8 1.628e3 r3 5 99 2.487e3 r4 6 99 2.487e3 c1 5 6 5.333e-12 eos 4 3 poly(1) (16,39) 0.25e-3 50.118 ios 2 3 5e-9 gb1 2 98 (24,98) 100e-9 gb2 4 98 (24,98) 100e-9 cin 2 3 1e-12 * * gain stage and dominant pole at 16 hz * eref 98 0 (39,0) 1 g1 98 9 (5,6) 402.124e-6 r7 9 98 497.359e6 c2 9 98 20e-12 v1 99 10 0.58 v2 11 50 0.47 d5 9 10 dx d6 11 9 dx * * common-mode stage with zero at 10 khz * ecm 15 98 poly(2) (3,39) (2,39) 0 0.5 0.5 r9 15 16 1e6
op179/op279 C16C rev. g outline dimensions dimensions shown in inches and (mm). c00290C0C1/02(g) printed in u.s.a. 8-lead tssop (ru-8) 8 5 4 1 0.122 (3.10) 0.114 (2.90) 0.256 (6.50) 0.246 (6.25) 0.177 (4.50) 0.169 (4.30) pin 1 0.0256 (0.65) bsc seating plane 0.006 (0.15) 0.002 (0.05) 0.0118 (0.30) 0.0075 (0.19) 0.0433 (1.10) max 0.0079 (0.20) 0.0035 (0.090) 0.028 (0.70) 0.020 (0.50) 8 0 8-lead narrow-body so (so-8) 0.1968 (5.00) 0.1890 (4.80) 85 4 1 0.2440 (6.20) 0.2284 (5.80) pin 1 0.1574 (4.00) 0.1497 (3.80) 0.0688 (1.75) 0.0532 (1.35) seating plane 0.0098 (0.25) 0.0040 (0.10) 0.0192 (0.49) 0.0138 (0.35) 0.0500 (1.27) bsc 0.0098 (0.25) 0.0075 (0.19) 0.0500 (1.27) 0.0160 (0.41) 8 0 0.0196 (0.50) 0.0099 (0.25) x 45 5-lead sot-23 (rt-5) 0.0079 (0.200) 0.0035 (0.090) 0.0236 (0.600) 0.0039 (0.100) 10 0 0.0197 (0.500) 0.0118 (0.300) 0.0590 (0.150) 0.0000 (0.000) 0.0512 (1.300) 0.0354 (0.900) seating plane 0.0571 (1.450) 0.0354 (0.900) 0.1220 (3.100) 0.1063 (2.700) pin 1 0.0709 (1.800) 0.0590 (1.500) 0.1181 (3.000) 0.0984 (2.500) 1 3 4 5 0.0748 (1.900) ref 0.0374 (0.950) ref 2 note: package outline inclusive as solder plating. revision history location page data sheet changed from rev. f to rev. g. edits to general description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 edits to pin connections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 edits to ordering guide . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 edits to absolute maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 edits to package types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 edits to outline dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

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