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july 2009 doc id 5656 rev 10 1/27 AN993 application note electronic ballast with pfc using l6574 and l6561 introduction dedicated ics for lamp ballast applications are now replacing the old solutions based on bipolar transistors driven by a saturable pulse transformer. the l6574 is a high-performance ballast driver, designed using 600-v bcd offline technology, which ensures all the features needed to properly drive and control a fluorescent bulb. it is provided with built-in vco start-up sequence circuitry, protections, and an operation amplifier for implementing a closed-loop control of the lamp current. www.st.com
contents AN993 2/27 doc id 5656 rev 10 contents 1 half bridge converter for el ectronic lamp ballast . . . . . . . . . . . . . . . . . . 4 1.1 lamp requirements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 2 l6574 ballast driver . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 3 device block description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 3.1 preheating and ignition section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 3.2 control section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 3.3 bootstrap section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 3.3.1 c boot selection and charging . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 4 description of the dem onstration application . . . . . . . . . . . . . . . . . . . 15 4.1 power factor section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 4.2 ballast section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 4.3 preheating and ignition sequence . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 4.4 current feedback loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 4.5 start-up and supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 4.6 safety circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 5 design tips . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 5.1 inductance and capacitor evaluation . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 6 dimming the lamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 6.1 dimming level and lamp turn-on . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 7 revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
AN993 list of figures doc id 5656 rev 10 3/27 list of figures figure 1. half bridge topology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 figure 2. internal block diagram of the l6574 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 figure 3. connection of a typical application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 figure 4. startup timing diagram and en2 function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 figure 5. timing block . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 figure 6. timing oscillator block . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 figure 7. cpre voltage and frequency shifting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 figure 8. operating frequency at cf = 470 pf . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 figure 9. controls timing diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 figure 10. startup timing diagram and en2 function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 figure 11. external bootstrap diode connection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 figure 12. l6574 integrated bootstrap diode connection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 figure 13. demonstration application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 figure 14. pcb and components layouts . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 figure 15. current feedback loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 figure 16. cpre waveform (ch1) and amplifier output (ch2) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 figure 17. open load safety circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 figure 18. extra voltage safety circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 figure 19. simplified schematic of the lamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 figure 20. preheating transfer function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 figure 21. operating transfer function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 figure 22. iterative process . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
half bridge converter for electronic lamp ballast AN993 4/27 doc id 5656 rev 10 1 half bridge converter for electronic lamp ballast voltage-fed, series-resonant half-bridge inverters are currently used for fluorescent lamps ( figure 1 ). this topology facilitate s operation in zero voltage switching (zvs) resonant mode, dramatically reducing the transistor switching losses and the electromagnetic interference. to design a cost-effective, compact and smart electronic lamp ballast, a dedicated ic could be used to drive directly the power mosfets of the half bridge. such controllers require a high voltage capability for the high -side floating transistor driver. 1.1 lamp requirements to prolong lamp life and to ensure efficient ignition of the lamp, the cathodes must be preheated. in fact, the preheating of the filaments allows an easy strike of the lamp, reducing the ignition voltage. during the preheating time, the lamp is characterized by a high impedance and the current flows only in the filaments. the resistance value of the filaments strictly depends on the type of lamp. typically, these filaments present an initial low value (a few ohms) that will increase by four to five times during the preheating phase. after the preheating phase, the lamp must be ignited by increasing the voltage across it. the ignition voltage value also depends on the type of lamp, and it increases with the aging of the lamp. for a typical tl 58 w, the ignition voltage value is not much less than 1000 v. when a simple inverter with a constant switching frequency is used, external circuitry is also necessary (for example, a ptc or discrete timer). however, with st?s l6574 smart controller, both the preheating and ignition functions are achieved by using simple resistors and a capacitor, which set all the start-up procedures. figure 1. half bridge topology am01 3 09v1 driver l res hv c res
AN993 l6574 ballast driver doc id 5656 rev 10 5/27 2 l6574 ballast driver the l6574, whose internal block diagram is shown in figure 2 , is an ic intended to drive two power mosfets or igbts in half-bridge topology, ensuring all the features needed to properly drive and control a fluorescent bulb. moreover, by varying the switching frequency, it is possible to modulate the current in the lamp and as a consequence, the output power as well. the device is available in dip16 and so16n packages. the l6574 has the following distinctive features. high voltage rail up to 600 v dv/dt immunity 50 v/ns in full temperature range driver current capability (250 ma source and 450 ma sink) switching times 80/40 ns rise fall with 1 nf load cmos shutdown input under-voltage lock-out preheat and frequency shifting timing sense operational amplifier for closed-loop control or protection features high-accuracy current- controlled oscillator integrated bootstrap diode clamping on vs so16, dip16 package. figure 2. internal block diagram of the l6574 am01 3 10v1 opout gnd v ref imin rign vco en1 v the v the en2 vs vboot out lvg uv detection v s hvg 5 6 7 4 2 3 1 12 9 8 10 11 14 15 16 bootstrap driver hvg driver lvg driver vthpre ifs cpre v ref imax rpre cf op amp + - opin- opin+ dead time driving logic control logic + - ipre + - + - + - level shifter cboot load h.v. bus
l6574 ballast driver AN993 6/27 doc id 5656 rev 10 figure 3. connection of a typical application am01 3 11v1 cpre cf rcs l_ballast c1 cres lamp d3 r2 rign. 2 7 6 5 1 9 8 11 14 15 16 12 3 410 l6574 rpre. + hvbus r3 r1 d4 r9 r7 c4 r8 c3 r6 r5 r4 csupply cboot cb qh ql csnub ref. d1 d2 rgl rgh c2 table 1. description of device pins number name function 1 c pre preheat timing capacitor. the capacitor c pre sets the preheating and the frequency shift time, according to the relations: t pre = k pre c pre and t sh = k fs c pre (typ. k pre = 1.5 s/ f, k fs = 0.15s/ f). this feature is obtained by charging c pre with two different currents. during t pre , this current is independent of the external components, so c pre is charged up to 3.5 v (preheat timing comparator threshold). during t sh , the current depends on the value of r pre (that is, on the difference between f pre and f ign ). in this way t sh is always set at 0.1t pre . in steady state the volt age at pin 1 is 5 v (see figure 5 ). 2 r pre maximum oscillation frequency setting. the resistance connected between this pin and ground sets the f pre value, fixing the difference between f pre and f ign (f pre > f ign ). at the end of the start-up procedure, the effect current drown from r pre is over. the voltage at this pin is fixed at v ref =2 v. 3 cf oscillator frequency setting. the capacitor c f , along with to r pre and r ign , sets f pre and f ing . in normal operation this pin shows a triangular wave. 4 r ign minimum oscillation frequency setting. t he resistance connected between this pin and ground sets the f ign value. the voltage at this pin is fixed at v ref =2 v. 5 opout out of the operational amplifier. to implem ent a feedback control loop this pin can be connected to the r ign pin by means of appropriate circuitry. 6 opin- inverting input of the operational amplifier. 7 opin+ non-inverting input of the operational amplifier.
AN993 l6574 ballast driver doc id 5656 rev 10 7/27 8 en1 enable 1. this pin (active high), forces the device into a latched shutdown state (like in undervoltage conditions). there are two ways of resuming normal operation: the first is by reducing the supply voltage below the undervoltage threshold and then increasing it again until the valid supply is recognized; the second is by activating the en2 input (see figure 9 ). the enable 1 is specifically designed for strong faults (for example, in case of lamp disconnection). 9 en2 enable 2. en2 input (active high) restar ts the start-up procedure (preheating and ignition sequence). this feature is useful if the lamp does not turn on after the first ignition sequence (see figure 10 ). 10 gnd ground. 11 lvg low-side driver output. this pin must be connected to the low-side power mosfet gate of the half bridge. a resistor connected between this pin and the power mosfet gate can be used to reduce the peak current. 12 vs supply voltage. this pin is connected to the supply filter capac itor (15.6 v typical). 13 n.c. not connected. this pin sets a distance be tween the pins related to the high-voltage side and those related to the low-voltage side. 14 out high-side driver floating reference. this pi n must be connected close to the source of the high-side power mosfet or igbt. 15 hvg high-side driver output. this pin must be connected to the high-side power mosfet gate of the half bridge. a resistor connected between this pin and the power mosfet gate can be used to reduce the peak current. 16 vboot bootstrapped supply voltage. the bootstr ap capacitor must be connected between this pin and out. a patented integrated circuit replaces the external bootstrap diode by means of a high-voltage dmos, synchronously driven with the low-side power mosfet. table 1. description of device pins (continued) number name function
device block description AN993 8/27 doc id 5656 rev 10 3 device block description the preheating control section and the bootstrap section are tightly linked to the application?s design. this chapter describes their workings and usage. 3.1 preheating and ignition section the l6574?s turn-on sequence is divided into three phases: the preheating phase, the ignition phase and the normal operation phase ( figure 4 ). the preheating phase is characterized by the highes t oscillation frequency (f max ) for a period t pre . during the ignition phase, the frequency shifts from f max to f min (which is the normal operating frequency) in a period t sh . all the above-mentioned parameters are set by carefully selecting a few external components. t pre and t sh are set by means of the capacitor c pre that is connected to pin 1. during the preheating phase (t pre ) the capacitor c pre is charged by means of a constant current i pre, which is generated internally and does not depend on any external components. the voltage across c pre increases linearly up to the "preheating threshold" at which the preheating phase terminates. equation 1 figure 4. startup timing diagram and en2 function am01 3 12v1 v(cpre) osc. freq. ignition steady state operation f max f min preheating t pre vsupply time il power-o.k. t pre k = c pre ? t ign 0.1 = t pre ?
AN993 device block description doc id 5656 rev 10 9/27 that is to say: equation 2 after the preheating time, the capacitor c pre is first quickly discharged and then recharged by the current i fs , generating a second voltage ramp which feeds a transconductance amplifier, as shown in figure 6 (the switch is closed). thus, this voltage signal is converted into a growing current which is subtracted from i max to produce the frequency shifting from f max to f min . the current that driv es the oscillator to set the fre quency during this shifting is equal to: equation 3 figure 5. timing block figure 6. timing oscillator block am01 3 1 3 v1 cpre ifs ipre discharge am01 3 14v1 cpre imin imax timing c s o m g iosc
device block description AN993 10/27 doc id 5656 rev 10 where: equation 4 r ign and r pre are the resistors connected to pin 4 and pin 2. at the end of the preheating time (t = t pre ), the l6574 oscillates at f max , set by: equation 5 this means that the preheating frequency depends on both r pre and r ign . at the end of the frequency shifting (t = t pre + t sh ), the second term of equation 3 decreases to zero and the switching frequency is set only by i min (that is, r ign ). equation 6 since the second term of equation 3 is equal to zero, we have: equation 7 note that there is no fixed voltage threshold across c pre in which the ignition phase finishes (that is, the end of the frequency shifting): t sh depends on c pre , i max , g m , and i fs ( equation 7 ). this fact is also verified in figure 7 . by making t sh independent of i max , the i fs current has been designed to be a fraction of i max , therefore: equation 8 in this way, the frequency shifting time depends only on the capacitor c pre . the typical value of the kfs constant (frequency shift ti ming constant) is 0.15 s/f, that is: equation 9 so when choosing c pre , both t pre and t sh are set. the frequencies f min and f max depend on the resistors r pre and r ign , but also on the capacitor c f (oscillator frequency setting -> capacitor at pin 3). f min is set by choosing c f and r ign , then with r pre the f = f max - f min is set. simplified equations can be used.
AN993 device block description doc id 5656 rev 10 11/27 equation 10 equation 11 these equations fit well with the measured values, especially in the frequency range of 30 to 100 khz. figure 8 shows a comparison between the measured and computed data of f min (at c f = 470 pf). to summarize: if r pre is not connected at pin 2, there is no f, so f max and f min have the same value. furthermore, there is no i max , that is to say there is no ifs current. in this case, c pre is charged only once by i pre = 2.3 a up to 3.5 v ( figure 7 shows the first voltage rise only). when c pre is discharged, no ifs current is present to recharge it. if r pre is connected after the preheating and ignition phases, pin 1(c pre ) is at 4.8 to 5 v with a current capability of a few a (1/6 ir pre ). if r pre is not connected, c pre is at gnd level in the normal operation phase. figure 7. c pre voltage and frequency shifting figure 8. operating frequency at c f = 470 pf table 2. parameter dependencies parameter set by choosing... t pre c pre t sh c pre f = f max - f max r pre f min r ign and c f am01 3 15v1 time time freq v(cpre) ignition preheating am01 3 16v1 40 60 80 100 120 r ign (k ) 20 40 60 80 100 120 f min (khz) measured values computed values
device block description AN993 12/27 doc id 5656 rev 10 3.2 control section the l6574 has two control functions, en1 and en 2. both functions are active high. to fully understand how these functions work, refer to figure 9 and figure 10 . en1 (latched enable) is dedicated to stopping all device functions and is usually activated when a strong fault is detected (for example, when the lamp is disconnected). this status can be cancelled in two different ways: either by reducing the supply voltage below its minimum threshold, or by activating the second control pin en2 (en2 overrides en1). when the device restarts , the start-up sequence is reactivated. en2 (repeating startup) is dedicated to restarting the preheating and ignition procedure; it is normally activated when the application fails to ignite the lamp. when en2 is activated the starting procedure restarts. the l6574 operates to the maximum frequency all the while that en2 remains active. figure 9. controls timing diagram am01 3 1 8 v1 lvg en1 vsupply time forced restart hvg latched enable en2
AN993 device block description doc id 5656 rev 10 13/27 3.3 bootstrap section bootstrap circuitry is needed to supply the high-voltage section. this function is normally accomplished by a high-voltage fast-recovery diode ( figure 11 ). in the l6574, a patented integrated structure replaces the external diod e. it is realized by means of a high-voltage dmos, driven synchronously with the low-side driver (lvg), with a diode connected in series, as shown in figure 12 . an internal charge pump ( figure 12 ) provides the dmos driving voltage. the diode connected in series to the dmos has been added to avoid it being unintentionally turned on. 3.3.1 c boot selection and charging to choose the proper c boot value, the external mos can be seen as an equivalent capacitor. this capacitor c ext is related to the total gate charge of the mos. equation 12 the ratio between the capacitors c ext and c boot is proportional to the cyclical voltage loss. it has to be: equation 13 for example, if q gate is 30 nc and v gate is 10 v, c ext is 3 nf. with c boot = 100 nf, the drop would be 300 mv. figure 10. startup timing diagram and en2 function am01 3 17v1 v(cpre) en2 t pre vsupply time ignition f max f min preheating il ignition foult-ignition preheating t ign (t sh )
device block description AN993 14/27 doc id 5656 rev 10 if the hvg needs to be supplied for a long period of time, the c boot selection also has to take into account the leakage losses. the internal bootstrap driver has great advantages: the external fast-recovery diode can be avoided (it usually has great leakage current). this structure can only work if v out is close to gnd (or lower) and if in the meanwhile the lvg is on. the charging time (t charge ) of the c boot is the time in which both conditions are fu lfilled and it has to be long enough to charge the capacitor. the bootstrap driver introduces a voltage drop due to the dmos r ds(on) (typical value at 25c is 150 ). at low frequencies, this drop can be neglected, but must be taken into account when the frequency is increased. the following equation is useful to compute the drop on the bootstrap dmos. equation 14 where q gate is the gate charge of the external power mosfet, r ds(on) is the on resistance of the bootstrap dmos, and t charge is the charging time of the bootstrap capacitor. for example: using a power mosfet with a total gate charge of 30 nc, the drop on the bootstrap dmos is about 1 v if the t charge is 5 s. in fact: equation 15 v drop has to be taken into account when the voltage drop on c boot is calculated: if this drop is too high or if the charging time is insufficient, an external diode can be used. figure 11. external bootstrap diode connection figure 12. l6574 integrated bootstrap diode connection v drop i ch e arg r ds on () v drop q gate t ch e arg ----------------------- r ds on () = = am01 3 19v1 to load h.v. hvg lvg c boot d boot v boot v s v out am01 3 20v1 ex_d99in1056 hvg lvg to load h.v. c boot v s v boot v out
AN993 description of the demonstration application doc id 5656 rev 10 15/27 4 description of the demonstration application the design has been developed to drive a tl fluorescent lamp up to 58 w. it is composed of two sections: the pfc, using the l6561 controller, and the ballast, based on the l6574 (see figure 13 and figure 14 ). the application is provided with a current feedback that can be used to control the power (and, if necessary, the dimming function) by varying the switching frequency during normal burning of the lamp. the application is also provided with a safety circuitry that gets activated when an open load or faulty ignition of the lamp is detected. figure 13. demonstration application circuit am01 3 21v1 stp4nb50/ stp4nk50z c12 100nf c13 1 f c12 470pf rs_2 0.68 bridge b1 fuse ntc 5 l2_2.1mh c6 22 f 450v dz1 14v c7 4.7 f 25v c8 100nk d2 stth1l06 c17 100nf 250v c18 8.2nf 1600v lamp d4 1n4148 r13 1.5k p1 4.7k r16 100k r18 100k r21 22 r22 22 stp4nb50/ stp4nk50z 2 7 6 5 1 9 8 11 14 15 16 12 3 4 10 l6574 r12 82k r19 10k r15 10k c9 8.2nf 1 2 6 5 4 7 8 l6561/2 r1 750k r6 68k r4 120k r7 22 r11 9.53k r10 750k r17 47 d5 q1_stp6nb50/ stp5nk50z c1 330nf 400v c2 10nf c4 680nf c11 680pf 630v r5 120k r29 6.8k r28 3.9k c16 0.47 f r25 20k c15 330nf r27 750k r26 750k r24 390k r20 6.8k c5 100nf r9 750k d3 1n4148 rs_1 sense t1 1.2mh (e25*13*7) c10 4.7nf r3 10k 3 1n4148 d99in1064 r2 750k r8 10k c3 220nf d1 1n4148 r14 10 r23 1k q3 q2 c19 100nf 2 1 4 5
description of the demonstration application AN993 16/27 doc id 5656 rev 10 figure 14. pcb and components layouts am01 3 22v1 41mm 190mm component l a yer s older l a yer
AN993 description of the demonstration application doc id 5656 rev 10 17/27 4.1 power factor section even if the pfc stage is not strictly necessary for electronic ballast applications, in this design it has been introduced for the following reasons. the pfc stage is necessary if the ballast input power is higher than 25 w. this provides many benefits. the front-end stage of conventional off-line converters, typically made up of a full wave rectifier bridge with a capacitor filter, gets an unregulated dc bus from the ac mains. therefore, the inst antaneous line voltage will be below the voltage on the capacitor most of the time, which means that the rectifiers will only conduct fo r a small portion of each line?s half-cycle. the current drawn from the mains will then be a series of narrow pulses whose amplitude will be 5-10 times hig her than the resulting dc value. lots of drawbacks result from this: much higher peak and rms current drawn from the line, distortion of the ac line voltage, overcurrents in the neutral line of the 3-phase systems, all contributing to a poor utilization of the power system's energy capability. this energy capability can be measured in terms of either total harmonic distortion (thd), as norms provides for, or power factor (pf), intended as the ratio between the real power (the one transferred to the output) and the apparent power (rms line voltage times rms line current) drawn from the mains, which is mo re immediate. a tradit ional input stage with a capacitive filter has a low pf (0.5-0.7) and a high thd (> 10%). the new european norms and the international standard requirements have spurred the design of high-power factor ballasts and are starting to impose a limit on the input current harmonic content. for these reasons, power fa ctor correctors (pfc) are now being widely diffused in consumer and industrial lighting. with a high power factor switching pre- regulator, interposed between the input rectifier bridge and the bulk filter capacitor, the power factor is improved (by up to 0.99). the current capability is incr eased, while the bulk capacitor peak current and the ha rmonic disturbances are reduced. the l6561/2 is an ic intended to control pfc pre-regulators by using the transition mode technique and is optimized for lamp ballast applications. the operation is summarized below (for more information, see an966). the ac mains voltage, that can range from 85 v to 265 v, is rectified by a diode bridge and delivered to the boost converter. the boost converter consists of a boost inductor (t1), a controlled power switch (q1), a catch diode (d2), an output capacitor (c6) and, obviously, control circuitry (see figure 3 ). the pfc section has been designed to supply a 400 v dc and a power of 60 w. 4.2 ballast section the regulated voltage is delivered to the ballast section. the ballast is based on the high- performance l6574, an offline half bridge driver designed using 600-v bcd technology. it adds to the fully-integrated half -bridge driver topology a built -in voltage controlled oscillator (vco), a preheating start-up procedure and an operational amplifier dedicated to the feedback loop. to avoid cross-conduction of the power mosfets or igbts, the internal logic ensures a minimum dead time.
description of the demonstration application AN993 18/27 doc id 5656 rev 10 the load consists of a series resonant circui t (l2-c18) with the lamp connected across the capacitor (c18). this topology allows operation in zero volt age switching mode, to reduce the transistor switching losses and the electromagnetic interference generated by the output wiring of the lamp. the blocking capacitor (c17) allows a zero average lamp current. in steady state the voltage across these capacitors is as high as half the high voltage bus, approximately 200 v. 4.3 preheating and ignition sequence the turn-on sequence can be divided in three phases: preheating, ignition and normal lamp burning. the preheating of the lamp filaments is achieved by a high switching frequency f pre , about 60 khz, set by r pre = r12+ p1 + r13 and cf = c12, to ensure that a current flows in the filaments without lamp ignition. in fact, the initial voltage applied across the lamp is below the strike potential. the duration of the preheating period t pre i s s e t by t h e c a p a c i t o r c pre = c13. the choice of this time is strictly dependent on the type of lamp. in the application t pre has been set to 1.5 sec. the ignition sequence begins after t pre . the switching frequency decreases towards the resonance point (l2-c18), increasing the voltage across the lamp and causing the ignition. the time interval in which the frequency shifts, t sh , amounts to t sh = t pre /10 = 150 ms. at the end of t sh the frequency reaches 31 khz (r18-c12), and the current feedback loop is then activated. 4.4 current feedback loop the current control is achieved by varying the switching frequency of the vco. since controlling the average current in the lamp means controlling th e output power, it is quite easy to perform the control function. the operational amplifier compares the low-pass filtered half-bridge current with a reference, achieved by a portion of the voltage at pin 2 (v pin2 = 2 v). this set-point could be changed by the trimmer p 1 to perform the dimming function. the amplifier?s output is connected to the ring pin by d 4 and r 16 . the diode d 4 is necessary to prevent the switching frequency from decreasing to below the value set by r18. at start-up the voltage across r s2 ( figure 15 ) remains low until the la mp ignites. as such, the inverting input of the amplifier (pin 6) also stays low, while the non-inverting input (pin 7) is set to a constant voltage (set-point) by the divider r 12 , p 1 and r 13 . therefore, the amplifier?s output (pin 5) remains high (5 v) until the lamp ignites, and d 4 is off. in this condition, the l6574 oscillates at f pre . once the lamp strikes on (after t pre and t sh ), the average voltage across rs2 increases and the feedback can regulate the lamp?s current.
AN993 description of the demonstration application doc id 5656 rev 10 19/27 4.5 start-up and supply the start-up procedure is very important in an application that contains two different sections. the ballast section starts before the pfc, avoiding any extra voltage at the output of the pfc section, and so the l6561 dynamic ovp ac tivation (see an966). this behavior is guaranteed under all conditions because the vs turn-on threshold of the l6574 is lower than that of the l6561/2. at start-up, the l6574 is powered by the resistor (r 4 + r 5 ). this resistor must be chosen so as to ensure the "before start-up current" of both the l6561 and l6574. when the ballast section is running, the charge pump (c11, r14, r17, d3 and dz1) enables supply to both devices. r17 and c10 enable reduction of noise at vcc. figure 15. current feedback loop figure 16. cpre waveform (ch1) and amplifier output (ch2) am01 3 2 3 v1 q3 stp4nb50 rs_2 0.68 c9 8.2nf r16 100k r18 100k d4 1n4148 r19 10k 65 4 7+ rign - am01 3 24v1
description of the demonstration application AN993 20/27 doc id 5656 rev 10 4.6 safety circuitry in normal operation the inductive load ensures a zero voltage switching mode, but if the lamp is disconnected the switching loss es in the power mosfets will increase considerably. to prevent this occurrence, a safety circuitry has been designed. when the lamp is connected the en1 input (pin 8) of the l6574 is held close to ground by the series of r 29 , the lamp filament and r s2 . if the lamp is not present, en1 is pulled up to vs by r 24 , forcing the l6574 into a latched shutdown state. to resume normal operation, the ballast must be turned off, and then on again. a second alarm has been designed to protect the application against any extra voltages which can arise if the lamp does not strike after the ignition sequence, for instance if the lamp is old. a partition of this extra voltage is rectified and delivered to the en2 input (pin 9) of the l6574, restarting the start-up procedure (preheating and ignition sequences). figure 17. open load safety circuit figure 18. extra voltage safety circuit am01 3 25v1 r24 390k r25 20k r29 6.8k c19 8.2nf 1600v lamp c16 .47uf +v s en1 (pin8) d98in816a am01 3 26v1 c17 100nf 250v lamp r28 3.9k r27 750k r26 750k c18 8.2nf 1600v r20 6.8k en2 (pin9) d5 1n4148 c15 330nf l2 2.1mh r23 1k
AN993 design tips doc id 5656 rev 10 21/27 5 design tips 5.1 inductance and capacitor evaluation to design an application with the l6574, a preliminary evaluation of the components can be done by fixing the lamp type and its electrical characteristics only. this evaluation is an "iterative" process bec ause some assumptions have to be made that need to be checked at the end of the process. figure 19 shows a simplified schematic of the load. in figure 19 r is the lamp filament resistance, rr is the operating lamp equivalent resistance when the lamp is off or during preheating. rr is an open circuit. first of all, one has to evaluate a proper inductance value. l has to provide the right current value to the lamp when it is already ignited and is working ("choke" inductance). therefore, l depends on the current required by the lamp, that is to say on the lamp?s operating wattage and voltage, on the operating frequency f min , but also on the voltage across l and the lamp (v b ). the greater the v b variation, the greater the inductance required to give a constant current to the lamp. v b variations are due to the high- voltage bus variations and to the ripple on the half battery capacitor (c b ). the ripple depends on the size of c b and a proper hypothesis has to be done on it to estimate the variation of v b . taking into account all this data and hypothesis, one can assume that during an operating condition most of the current flows into the lamp, not into c, and all the power delivered to the system is delivered to the lamp. v lamp is the operating voltage across the lamp and p lamp is the operating lamp wattage, so a good approximation to conduct the choke inductance is: equation 16 the second step is the evaluation of the capacitance across the lamp (c). figure 19. simplified schematic of the lamp am01 3 27v1 cb c l lamp cb c l rr r r
design tips AN993 22/27 doc id 5656 rev 10 when the lamp is not yet ignited, c has to allow a sufficient current to flow into the lamp filament in order to preheat correctly. the power to be delivered to the lamp filaments (pfil), the preheating frequency, the lamp filament resistance r and the maximum voltage to be applied across c without causing lamp ignition are constraints that help to evaluate the size of the capacitor. by setting the current through the lamp filament and the maximum voltage across the capacitor, one obtains a range of capacitor values. equation 17 these l, c b and c values have to be corrected to obtain standard and commercial component values. using these values and lamp equivalent resistances, the transfer functions during the preheating and operating conditions can be calculated. the preheating transfer function allows you to see if, when moving towards the resonant frequency (l-c), there is a frequency at which the voltage across the lamp enables its ignition. this frequency has to be between f max and f min . the gain of the transfer function depends on r also, but r changes greatly during the preheating phase (also 3 to 4 times) and this must be taken into consideration (see figure 20 ). the operating transfer function can be used to check if the operating voltage across the lamp (at f = f min ) is similar to the one used to evaluate l ( figure 21 ). figure 20. preheating transfer function am01 3 2 8 v1 f=f(l,c,r@25c) v lamp v in freq f=f(l,c,r@tpre) f pre 1/2 ?? sqrt(l ? c) f min frequency shifting during t sh
AN993 design tips doc id 5656 rev 10 23/27 if one of these conditions is not verified, the evaluation process has to be redone changing the initial hypothesis (frequencies or assumption on the cb ripple). if everything concords, the values found can be used and a preliminary stage of design can be concluded. further experiments may be required and the initial assumptions may need to be verified to correctly set the components and frequency values, and if necessary another evaluation process reiterated with better assumptions. the following flow chart can help the iteration process. figure 21. operating transfer function figure 22. iterative process am01 3 29v1 f = f(l,c,rr) f min v lamp v in freq am01 33 0v1 l evaluation c evaluation data: operating lamp elec.characteristic hvb hypothesis: fmin cb hypothesis: fpre data : ignition and preheating lamp elec.characteristic transfer preheating funct. transfer operating funct. check on board ok ok ok
dimming the lamp AN993 24/27 doc id 5656 rev 10 6 dimming the lamp the lamp is dimmed by changing the working frequency. the minimum working frequency is: equation 18 to change the working frequency, one has to change the current that flows from pin 4. using the circuitry shown in figure 15 , the current that flows from pin 4 also depends on the amplifier?s out pin (pin 5) voltage: if v5 > v4, there will be no current in r 16 and the frequency will be equal to f min . on the other hand, when the amplifier?s out voltage goes lower than v4, the current that flows from pin 4 will go into r 16 and r 18 . pin 4 sees the parallel between r 18 and an equivalent resistor that depends on v 5 and on the voltage of d 4 . equation 19 so the working frequency will be: equation 20 and the max f working is obtained when v5 = 0 v. for example, we can calculate a maximum frequency of ~53 khz (v4 = 2 v and assuming vdiode3 = 0.5 v). the dimming level is set by changing the value of the p1 potentiometer, which causes the amplifier?s positive reference to change from 20-30 mv to 110-120 mv. if r16 is lowered, the maximum working frequency is increased and the dimming level lowered as well (the higher the frequency, the lower the current in the lamp?s arc). if one tries to dim the lamp towards a low power range (<20% of the electrical arc power) a common effect is the presence of stationary waves along the lamp tube and/or some flickering effects. a common trick to get rid of these disturbances is to add a small continuous current flow inside the lamp (a few ma). the easiest way to do this is to add a resistor in parallel to the half battery capacitor (c17 in figure 13 and figure 18 ). for instance, one can add 50 k (i = 200 v/50 k = 4 ma), which is already very effective. of course, the resistor has to be able to sustain the power dissipation caused by the current flow, so it is common to use many resistors connected in parallel. remember that this is only a tip. it helps but is not enough, and should be used together with the right frequency settings that must be chosen according to the type of lamp. 4 4
AN993 dimming the lamp doc id 5656 rev 10 25/27 6.1 dimming level and lamp turn-on during the start-up sequence the frequency always goes from f max to f min , regardless of the dimming level, and it is only after this that the lamp?s turn-on frequency moves towards higher frequencies. the delay during which f = f min allows the lamp to turn on, but has the drawback of causing a "flash" that can be unpleasant. as the minimum length of this delay depends on the type of lamp, it is better to set it as low as possible, finding the best compromise. the easiest way to set this delay is to act on the amplifier?s compensation, that is, on c9: the higher c9, the longer the delay and the easier the lamp turn-on. values up to 200 nf and more are commonly used.
revision history AN993 26/27 doc id 5656 rev 10 7 revision history table 3. document revision history date revision changes 21-jun-2004 9 changed: figure 12 to 17 and changed formula on the page 19 13-jul-2009 10 document reformatted. english reviewed. modified: equation 19 and section 6 .
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